Gerhard Mercator Universität Gesamthochschule …Gerhard-Mercator-Universität Gesamthochschule...

Gerhard MercatorUniversitätGesamthochschuleDuisburg

Annual Report 1996/97

Department ofOptoelectronics

Gerhard-Mercator-UniversitätGesamthochschule Duisburg

Fachbereich ElektrotechnikFachgebiet Optoelektronik

ZHOLotharstr. 55

D - 47057 DuisburgGermany

Prof. Dr. rer. nat. D. Jäger

+49-203-379-2340

+49-203-379-2409

[email protected]

http://www.oe.uni-duisburg.de

Editor: R. Buß

Head:

Tel:

Fax:

Email:

URL:

Pictures of the Center for Solid-State Electronics and Optoelectronics taken in Dec. 1997

4

The Center for Solid-State Electronics and Optoelectronis (ZHO)

After only nine month of construction the topping-out-ceremony of the new Center for Solid-State

Electronics and Optoelectronics (Zentrum für Halbleitertechnik und Optoelektronik, ZHO) has been

celebrated on June, 4th, 1997.

The center is planned to be the new home of the Department of Optoelectronics and the Solid-

State Electronics Department in the beginning of October 1998. It consitst of two parts: The clean-

room building with an area of approx. 470 m2 and the building for the offices and laboratories with an

area of approx. 1200 m2.

Table of Contents

1 Foreword

2 Members of the Department

3 Research

3.1 Optical Networks3.1.1 High-speed, high-power travelling-wave photodetectors

3.1.2 Fabrication and characterization of a travelling-wave photodetector

3.1.3 Simulation of the microwave generation of a travelling-wave photodetector

3.1.4 Determination of RF-equivalent circuit elements of travelling-wave photode-

tectors using network analysis

3.1.5 Polarization insensitive waveguide modulators on InP

3.2 Optical Interconnects and Processors3.2.1 Neurotechnology: Retina Implant

3.2.2 Analysis of the optical energy and signal transfer module for an artificial vision

prosthesis

3.2.3 Development of an optical signal and energy transmission system

3.2.4 Infrared data link for rotating display-systems

3.2.5 Nonlinear hybrid GaAs/AlGaAs multilayer-heterostructures for high-speed

information processing

3.2.6 8 x 8 LED arrays integrated with 64 channel Si-driver circuits

3.3 Millimeterwave Electronics3.3.1 Picosecond pulse generation on monolithic nonlinear transmission lines using

high-speed InP-HFET diodes

3.3.2 Millimeter wave power generation nonlinear transmission lines

3.3.3 Nonlinear RTD circuits for high-speed A/D conversion

3.4 Optical Sensor Systems3.4.1 MQW-Electroabsorption-Modulator for application in a fiberoptic fieldsensor

3.4.2 Photovoltaic cells for fiber optic EMC - Sensor power supply

3.4.3 Time- and frequency-domain electro-optic field mapping of nonlinear trans-

mission lines

7

9

11

1111

14

17

20

25

2929

32

35

38

41

48

5151

55

59

6262

64

66

5

3.4.4 Characterization of monolithic microwave integrated circuits by heterodyne

electro-optic sampling

3.4.5 Development of an experimental setup for field probe measurements on

nonlinear transmission lines

3.5 Technologies for Optoelectronic Components and Systems3.5.1 Development of a measurement system for the optical characterization of

full-colour-LED-displays

3.5.2 Opinion poll on the evaluation of the legibility of LED-based displays

3.5.3 Evaluation of possible improvements to enhance the UV-power efficiency of

a xenon flashlamp system

3.5.4 Construction of a flip chip device for bonding integrated circuits

4 Teaching activities

4.1 Lectures, excercises, and practical studies

4.2 Seminars and colloquia

4.3 Doctoral, Diploma, and Graduate theses

5 Publications and presentations

6 Guide to the Department of Optoelectronics

TABLE OF CONTENTS6

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95

100

103

107

1 Foreword

The Fachgebiet Optoelektronik at the Ger-

hard-Mercator-Universität Duisburg has a tradi-

tion of research and teaching excellence dating

back to its establishment in 1989/90. This Re-

port provides a summary of major research in-

volvements and teaching activities reviewing

also publications and presentations by the mem-

bers of the institute .

The last two years were characterized by a

further-broadening of our scientific networks and

additional projects funded by various external

institutions. A first key research area is micro-

wave photonics with special emphasis on trav-

elling-wave photodetectors, electro-absorption

modulators and nonlinear optoelectronic devic-

es, where system aspects played a continuous-

ly increasing role. Special emphasis has further

been laid upon the two-dimensional electro-op-

tical characterization of monolithic microwave

integrated circuits and high-speed devices. As

a third topic, nonlinear optics and optoelectron-

ics in III-V-heterostructures and pulse compres-

sion in nonlinear MMICs were studied in detail.

Additionally to our activities within the Sonder-

forschungsbereich 254, mayor funding has been

provided by the Retina Implant project (EPI-

RET) and the collaborative programme on the

development of an EMC-field sensor, where our

Fachgebiet acts as the coordinator. We are al-

ready proud of looking back to a relevant exhi-

bition during the Laser 97 Fair in Munich. An-

other remarkable event was the International

Topical Meeting on Microwave Photonics

(MWP) held at the moated castle Schloß Hu-

genpoet in September 1997 which was orga-

nized by our Fachgebiet, D. Jäger being simul-

taneously the Chair of the International Steering

Committee on MWP (details on next page).

7

As a result of their remarkable research work,

Dr.-Ing. G. David received a fellowship of the

Alexander von Humboldt-Stiftung, funding a two-

years stay at the University of Michigan. More-

over, Dr.-Ing. A. Stöhr was awarded a grant to

carry out research work at the Communications

Research Laboratories, Ministry of Posts & Tele-

communications in Tokyo. Further, D. Jäger re-

ceived the title Professor Onorific from the

University of Brasov/Romania and became the

Chair of the German IEEE/LEOS Chapter.

Besides the usual and obligatory courses, the

Fachgebiet Optoelektronik offered in 1997 a new

lecture Einführung in die Multimediatechnik -

Technologien, Systeme, Anwendungen. More-

over, the Institute was involved in the Duisburg

Summerschool for Women, the Tag der Fors-

chung and the organisation of research activi-

ties on photonic bandgap materials in the frame-

work of the Forum Materialforschung in our

university. Finally, we note with great pleasure

that T. Alder and D. Kalinowski have been

awarded the University Price for excellent diplo-

ma theses.

I wish to thank all friends inside and outside

the university for their continuous encourage-

ment and assistance. Also, I would like to ex-

press my sincere thanks to all members of the

Institute for their efforts and contributions to our

success in optoelectronics.

Duisburg, September 1998

8

he 1997 International Topical Meeting

on Microwave Photonics (MWP97)

has been held from September 3 through 5

in the historical buildings of the majestic 17th

century moated Castle Schloß Hugenpoet,

situated in the beautiful Ruhrtal countryside

in the south of Essen near Duisburg.

This 7th Topical Meeting in the series on this

subject followed those in Cernay-la-Ville, France

(1994), Keystone, U.S.A. (1995), and Kyoto,

Japan (1996). It was the first one held in Ger-

many and has been organized by the Gerhard-

Mercator-Universität-Duisburg. On September

3, the Meeting started with a Workshop entitled

Photonic technologies for phased array anten-

nas where 6 invited papers addressed recent

results in this continously growing field of re-

search. In the Plenary Session on September 4,

3 invited speakers, Dr. R. Heidemann, Alcatel

SEL AG, Stuttgart, Germany, Dr. M.J. Wale,

GEC-Marconi Materials Technology Ltd,

Northampton, U.K., and Dr. D. Novak, The Uni-

versity of Melbourne, Australia, presented lec-

tures on the topic Microwave Photonics:

Present and Future. The regular conference

program consisted of 7 sessions on topics such

as Optical generation of microwave signals,

Optoelectronic modulators, mixers, and receiv-

ers, Microwave photonic systems, Fibre radio

networks, Modelling in microwave

photonics, and Microwave photonics

for measurements. Each session has

been opened by invited senior tech-

nologists from France, Hong Kong,

Japan, U.K., U.S.A., and Germany

having provided additional impetus to

this multi-disciplinary research area

of microwave photonics.

The whole program included fur-

theron a video session via internet with KDD in

Tokyo, Japan, a poster session and was com-

pleted by a postdeadline session. More than 100

papers have been submitted from 13 countries

showing the increasing interest of scientists and

engineers in this area. After careful evaluation

the Technical Program Committee has recom-

mended 69 papers - 54 oral and 15 poster - for

presentation. Additionally, 4 papers have been

selected during the conference for the postdead-

line session. 140 scientists and engineers have

registered for the Topical Meeting. In addition

to the technical program, a Partners Program,

a Welcome/Barbecue Party and a Gala Dinner

have been organized.

The Meeting has been sponsored by the

Deutsche Forschungsgemeinschaft (Bonn, Ger-

many), Hewlett-Packard GmbH (Ratingen, Ger-

many), Institut für Mobil- und Satellitenfunktech-

nik GmbH (Kamp-Lintfort, Germany), Lucent

Technologies (Allentown, P.A., U.S.A.), and the

Gerhard-Mercator-Universität Duisburg. More-

over it has been cooperatively sponsored by the

IEEE MTT-S and LEOS including the German

Chapters.

The organizers of MWP97 look back to a very

fruitful conference and look forward to the up-

coming meetings MWP98 in Princeton, N.J.,

U.S.A., and MWP99 in Melbourne, Australia.

T

2 Members of the Department

Department of OptoelectronicsZHO, Lotharstr. 55

47057 Duisburg, Germany

fon: +49 203 379-2340

fax: +49 203 379-2409

Head of the DepartmentJäger, Dieter Prof. Dr. rer. nat.

SecretaryGappa, Ulrike Optoelectronics

Tempel, Karin SFB 254

ScientistsAlles, Martin Dipl.-Ing.

Alder, Thomas Dipl. Ing.

Braasch, Thorsten Dipl.-Phys.

Buß, Rüdiger Dipl.-Ing.

David, Gerhard Dr.-Ing.

Groß, Matthias Dipl.-Phys.

Heinzelmann, Robert Dipl.-Ing.

Hülsewede, Ralf Dipl.-Phys.

Jäger, Irina Ph. D.

Kalinowski, Dirk Dipl.-Ing.

Knigge, Steffen Dr.-Ing.

Kremer, Ralf Dr.-Ing.

Redlich, Stefan Dipl.-Ing.

Schmidt, Manuel Dipl.-Phys.

Stöhr, Andreas Dr.-Ing.

Wingen, Georg Dipl.-Phys.

Zumkley, Stefan Dr. rer. nat.

Guest ScientistsDragoman, Mircea Prof. Dr.

Johnson, Roger Dipl.-Ing.

Lee, Chi Prof. Dr.

Mezentsev, Vladimir Ph. D.

Wendrix, Veronique Dipl.-Ing.

TechniciansMang, Sabine

Schedwill, Veronique

Slomka, Heinz Ing. grad.

StudentsAppenrodt, Nils

Balci, Senay

Baumeister, Thomas

Berger, Oliver

Boscher, Guido

Brings, Ludger

Bussek, Peter

Christoffers, Niels

Einweck, Michaela

Engel, Thomas

Ervens, Jutta

Hedtke, Ralph

Heinzdorf, Michael

Jabs, Mirco

Kampermann, Claus

Kreuder, Andreas

Lüdeke, André

Lotz, Oliver

Manh-Duc, Ngo

Meininger, Mark

Moeck, Jens-Peter

Neuhaus, Birgit

Ponellis, Bernd

Reintjes, Stefanie

Rogall, Michael

Spiegeler, Britta

Wenning, Michael

Weimann, Uwe

9

3 RESEARCH10

3 Research

3.1 Optical Networks

3.1.1 High-speed, high-power travel-ling-wave photodetectors

M. ALLES

ecently, new communication sys-

tems combining the advantages of

wireless transmission and fiber optics have

been proposed. Applications are transmis-

sion of traffic information, multimedia, or In-

ternet access. These systems operate

usually at an optical wavelength of 1.55µm

at frequencies up to 60GHz. Since the pho-

todetector should generate as much electri-

cal power as possible, the travelling-wave

photodetector under investigation has to ful-

fill these requirements.

IntroductionNovel wireless millimeterwave communica-

tion systems have been proposed by several

groups, see for example [1-4]. These systems

use fiber optics to transmit the signals over large

distances. Fiber optic cables have very low at-

tenuation of about 0.8dB per km, which is also

independent from the signal frequency, where-

as loss of coaxial cables is about 100dB per km.

An optical source generates a heterodyne

signal at 1.55µm, where the difference frequen-

cy of the two optical carriers corresponds to the

electrical millimeterwave frequency. The data

signal is modulated on one optical carrier. The

optical heterodyne signal is distributed to an

antenna station using usual fibre optic compo-

nents. At this antenna station, the photodetec-

tor converts the optical signal to a millimeter-

wave, which is amplified and transmitted to a

remote station. These communication systems

should be used for traffic information, digital vid-

eo, multimedia, or Internet access.

This system approach leads to some impor-

tant requirements for the photodetector. The

photodetector has to work at an optical wave-

length of standard fiber optics, i.e. at 1.3/1.55µm.

In a further point, the photodetector should be

capable to operate in the high frequency regime

to generate electrical signals in the millimeter-

wave regime. Additionally, the photodetector has

to generate as much electrical output power as

possible to reduce the requirements of the

electrical amplifier used in the antenna station.

Usually, high-speed photodetectors are fab-

ricated as lumped devices which are RC-time

limited. This means that high bandwidths can

only be reached if the device size is scaled down

to the micrometer regime. Dimensions of photo-

detectors operating at 100GHz are about 10µm2

[5]. Due to the small device size the photode-

tectors can only operate at low optical input pow-

ers to avoid saturation effects in the small vol-

ume. This limitation can be overcome if the

travelling-wave concept is considered for the

development of high-speed photodetectors [6].

High-speed travelling-wave photodetectorsThe travelling-wave photodetector is fabricat-

ed as an optical waveguide which is coupled to

an electrical waveguide due to an optical ab-

sorption layer. The capacitance of the electrical

waveguide is compensated by the inductance

of the transmission line resulting in an electrical

bandwidth which is not RC-time limited. This

concept avoids scaling down the device dimen-

sions. In contrast, the travelling-wave photode-

tector can be fabricated as a distributed device

in order to reduce optical saturation effects.

3.1 Optical Networks 11

R

Fig. 1: Sketch of the travelling-wave photodetector. Popt and Pel are the input optical and output electri-

cal powers,

The structure of the fabricated travelling-wave

photodetector is depicted in Fig. 1. The device

consists of an active region and a taper at the

end of the structure, used for hybrid integration

with electrical millimeterwave amplifiers. The

photodetector is MBE-grown on a semi-insulat-

ing InP-wafer for operation in the 1.3/1.55µm

regime. An InGaAlAs layer is used as an optical

waveguide. An InGaAs absorbing layer, leak-

age coupled to the waveguide, generates elec-

tron-hole pairs. Finally, an InAlAs-layer as a clad-

ding layer and an InGaAs/InGaAlAs superlattice

as a Schottky-barrier enhancement layer are

grown.

Electrical waveguiding is achieved using a

coplanar transmission line. The outer conduc-

tors form ohmic contacts to the n+ doped region

of the optical waveguide, whereas the center

conductor is fabricated as a Schottky contact.

The depletion layer underneath the center con-

ductor separates the photo-generated electron-

hole pairs in the absorption region.

Since the travelling-wave photodetector uses

the interaction of optical and electrical waves,

the optical and the electrical phase velocity

should be equal (phase-matching condition).

This can be achieved due to the fact, that the

Schottky-contact generates slow-wave effects

on the electrical transmission line [7].

The efficiency of the travelling-wave photo-

detector can be calculated numerically using a

distributed equivalent circuit model for genera-

tion and propagation of electrical waves on the

coplanar transmission lines (see Calculation of

the electrical millimeterwave generation of a trav-

elling-wave photodetector in this annual report).

A distributed current source describes the im-

pressed photocurrent per unit length due to elec-

tron-hole generation in the depletion layer. The

numerical calculation of the output power leads

3 RESEARCH12

to an electrical millimeterwave power of -

18.4dBm at 40GHz for a travelling-wave photo-

detector with an active length of 1mm. The opti-

cal wavelength is 1.3µm and the input power is

0dBm per optical carrier.

The electrical output power of the travelling-

wave photodetector has been measured using

an optical heterodyne setup. Two tunable 1.3µm

Nd:YAG-lasers generate an optical heterodyne

signal with beating frequencies in the millime-

terwave regime. The generated electrical sig-

nal is measured using a coplanar on-wafer probe

and a spectrum analyzer.

The fabricated travelling-wave photodetector

leads to an electrical output power of -19.7dBm

at a reverse bias voltage of 12V and a frequen-

cy of 40GHz using the heterodyne measurement

setup, which is in good agreement with the the-

oretically determined value.

Since the capacitance of this device is 1.2pF,

the resulting RC -time constant in a 50W sys-

tem would lead to a 3dB-frequency of about 2.5

GHz, which is far below the measured frequen-

cy. This shows the validity of the travelling-wave

concept to overcome RC-time limitation.

ConclusionFuture wireless millimeterwave communica-

tion systems using optical heterodyne tech-

niques are described. The requirements for high-

speed photodetectors are discussed. Since high

bandwidth and large electrical output power are

needed simultaneously, travelling-wave photo-

detectors are under investigation. Up to now,

an electrical output power of -19.7dBm at a fre-

quency of 40 GHz could be measured.

AcknowledgmentThe author would like to thank U. Auer (Fach-

gebiet Halbleitertechnik/-technologie) for grow-

ing the epilayer and for fabrication of the travel-

ling-wave photodetector.

References[1] R. Heidemann, R. Hofstetter, H. Schmuck,

60GHz fibre-optic distribution technology for traf-

fic information and multimedia, IEEE, MTT-S

and LEOS Topical Meeting on Optical Microwave

Interactions, Proc. pp. 133-136, Abbaye des

Vaux de Cernay, 1994

[2] J. Park, K.Y. Lau, Millimetre-wave (39GHz) fi-

bre-wireless transmission of broadband multi-

channel compressed digital video, Electron.

Lett., pp. 474-476, Vol. 32, 1996

[3] D. Wake, C.R. Lima, P.A. Davies, Transmission

of 60-GHz signals over 100km of optical fibre

using a dual-mode semiconductor laser

sources, IEEE Photon. Techn. Lett., pp. 578-

580, Vol. 8, 1996

[4] E. Boch, High bandwidth mm-wave indoor lo-

cal area networks, Microwave Journal, pp. 152-

158, 1996

[5] K. Kato, A. Kozen, Y. Muramoto, Y. Itaya,

T. Nagatsuma, M. Yaita, 110-GHz, 50%-effi-

ciency mushroom-mesa waveguide p-i-n photo-

diode for a 1.55-µm wavelength, IEEE Photon.

Techn. Lett., pp. 719-721, vol. 6, 1994

[6] D. Jäger, Optical Information technology, ed.

S.D. Smith and R.F. Neale, Springer-Verlag, pp.

328-333, 1193

[7] D. Jäger, Slow-wave propagation along variable

Schottky contact microstrip line, IEEE Trans. Mi-

crowave Theory and Techn., pp. 566-573, vol.

24, 1976


InP

InGaAlAs : Si

InGaAs SQW/MQWInAlAs

Fig. 1: Travelling-wave photodetector layer structure.

3.1.2 Fabrication and characteriza-tion of a travelling-wave photodetec-tor

V. WENDRIX AND M. ALLES

ecently, high-speed travelling-wave

photodetectors are under investiga-

tion as optoelectronic power converters used

in future communication systems. In this

work the fabrication of high-speed travelling-

wave photodetectors is described. The fab-

ricated devices are characterized using

standard measurement techniques. The elec-

trical millimeterwave generation is deter-

mined using an optical heterodyne setup with

two 1.3µm Nd:YAG lasers.

IntroductionFor future communication systems combin-

ing fiber optic links with wireless transmission

techniques, high-speed photodetectors are

needed. These photodetectors should be used

for hybrid integration with millimeterwave am-

plifiers. A taper at the output end of the photo-

detector facilitates flipchip- or wire-bonding with

additional devices. The fabrication of travelling-

wave photodetectors with tapers is described in

this work [1].

Fabrication of travelling-wavephotodetectors

For the fabrication of a travelling-wave pho-

todetector several etch steps, a polyimide step

and metalization steps are needed [2]. The MBE-

grown wafers, which contain the necessary lay-

ers are depicted in Fig. 1 are fabricated in the

Department of Optoelectronics. The travelling-

wave photodetector contains in general three

different layers grown on a semi-insulating InP

wafer.

An InGaAlAs layer acts as optical waveguide,

an InGaAs quantum well layer provides optical

absorption, and, finally, an InAlAs layer is used

as a cladding layer. The metalization of the trav-

elling-wave photodetector is fabricated as an

electrical coplanar waveguide. The center con-

ductor forms a Schottky contact to the InAlAs -

layer. The outer metalization is evaporated on

the InGaAlAs-layer. This metalization is alloyed

in order to form ohmic contacts to the n-doped

semiconductor. The taper is fabricated at the

output end of the travelling-wave photodetec-

tor. In order to reduce millimeterwave attenua-

tion and for a characteristic impedance of 50W,the metalization of the taper has to be fabricat-

ed directly on the semi-insulating InP-wafer.

Therefore, an insulation of the edge of the me-

sas is necessary to prevent a short circuit be-

tween the center conductor and the outer met-

alization.

The processing of the travelling-

wave photodetector starts with two

etch steps. The etching defines the

lateral dimension of the optical

waveguide and of the absorbing lay-

er, cf. Fig. 2.

3 RESEARCH14

R

InP (b)(a)

Fig. 2: Etching of the two mesas, (a) cross-section, (b)

top-view.

InP (a) (b)

Fig. 3: Polyimide step, (a) cross-section, (b) top-view.

InP (a) (b)

Fig. 4: Evaporation of the metalization for the ohmic

contacts, (a) cross-section, (b) top-view.

InP (a) (b)

Fig. 5: Processing of center conductor and taper, (a)

cross-section, (b) top-view.

The fabrication of the mesa structure is done

using wet chemical etching with a liquid etchant

consisting of H3PO4:H2O2:H2O (1:1:40). This

etch system has almost no effect on InP.

The insulation of the mesa edge is done us-

ing a non-conducting polyimide step (Probimid

408), Fig. 3. The polyimide is processed

as a negative light sensitive resist and de-

veloped with a polyimide developer. A hard

bake process at 350°C makes the polyim-

ide resistant for the following process

steps.

In the next step, the metalization for the

ohmic contacts is evaporated on top of the

n-doped InGaAlAs-layer, Fig. 4. The met-

alization consists of Ge (30nm), Ni (5nm),

and Au (300nm). The metallic layers are

alloyed at about 550°C. This leads to ohm-

ic contacts with low impedances between

the metalization and the semiconductor.

The fabrication of the travelling-wave

photodetector finishes with the metaliza-

tion of the center conductor and the taper,

Fig. 5. Since the center conductor should

form a Schottky contact to the semicon-

ductor, it is necessary to evaporate Pt/Ti/

Pt/Au or Cr/Au.

MeasurementsThe characterization of the travelling-

wave photodetector is first done using cur-

rent-voltage and capacity voltage mea-

surements. The current-voltage

measurement gives information about the

functionality of the Schottky-contact diode

formed between the center conductor and

the outer metalization. As can be seen

from Fig. 6, the fabricated devices show

the typical current-voltage characteristic of a di-

ode.

In forward direction the current raises expo-

nentially with increasing voltage. With reversed

bias, the diode shows a high dark current which

increases with the voltage applied to the device.

The build-in voltage of this diode is about 0.6V.


U (V)-2 -1 0 1 2

0.001

0.01

0.1

1

10

100

Fig. 6: Current-voltage measurement of a travelling-

Spektrum-analyzer

lens DUT

on-waferprobe

Multi-meter

Laser 2

Laser 1

lens

Bias-Tee

Fig. 7: Heterodyne measurement setup using monomode fibers.

The capacitance-voltage measurements al-

lows the determination of the doping level of the

fabricated structures. The capacitance C of a

depletion layer is given by:

C AN

U UkT

q

r D

B

=− − −

ε ε0

2

with the area of the depletion layer A, the

dielectric constant er e0, the density of do-

nor dopants ND, the build-in voltage UB, the

applied bias voltage U, Boltzmanns con-

stant k, the temperature T, and the elec-

tron charge q.

If the capacity is known, it is possible to

calculate the density of donor dopants:

( )Nq A

U

CD

r

= − 2

102 2ε ε

d

d

The capacity-voltage measurement

leads to a doping level of about 1018cm-3

in the InGaAlAs-layer and less than

1017cm-3 in the InGaAs and the InAlAs-lay-

er.

Optical heterodyne setupFor the measurement of the optoelectronic

conversion efficiency, a heterodyne setup with

two Nd:YAG-lasers operating at 1.3µm is used.

The wavelength of the lasers is adjustable by

detuning the temperature of the laser head. Up

to now, the two lasers illuminate the travelling-

wave photodetector via free space and a micro-

scope lens. To facilitate

the optical coupling to the

photodetector the use of

optical fibers has been

investigated. This mea-

surement setup is shown

in Fig. 7.

Each laser is coupled

to a monomode fiber.

Both fibers are coupled

to a third monomode fi-

ber using a GRIN-lens.

Finally, this fiber is direct-

3 RESEARCH16

L’

R’

C’ I’

G’C’

Vdc

m

G’ C’L

bb

hl

R’

prlz rlz

Fig. 1: Equivalent circuit model of the travel-

ling-wave photodetector. All elements are per

unit length.

I

ly coupled to the device under test (DUT).

The electrical measurement of the optoelec-

tronically generated millimeterwave is achieved

using a coplanar on-wafer probe, a bias-tee to

separate the high-frequency and the dc-signals,

a multimeter for photocurrent, and an spectrum

analyzer to measure the amplitude of the milli-

meterwave in frequency domain.

ConclusionIn this report, the fabrication of travelling-wave

photodetectors is described. Photodetectors

have been processed successfully. Character-

ization of these devices is done using current-

voltage and capacity-voltage measurements.

Finally, the heterodyne measurement setup has

been investigated in order to facilitate optical

coupling to the photodetectors using monomode

fibers.

References[1] V. Wendrix, Fabricatie en karakterisatie van een

Traveling-Wave Photodetector, Diploma thesis,

Department Toegepaste Natuurkunde, Vrije

Universiteit Brussel in cooperation with

Fachgebiet Optoelektronik, Gerhard-Mercator-

Universität Duisburg, 1996

[2] R. Haupt Experimentelle Untersuchungen zur

Integration von Schottky-Kontakt-Varaktordioden

für den Einsatz in periodischen Leitungs-

strukturen, Graduate thesis, Fachgebiet

Optoelektronik, Gerhard.Mercator-Universität

Duisburg, 1995

3.1.3 Simulation of the microwavegeneration of a travelling-wave pho-todetector

A. LUEDEKE AND M. ALLES

n this report the photoelectronic micro-

wave generation of a coplanar In-

GaAlAs/InP Schottky-contact travelling-wave

photodetector (TWPD) is analyzed by numer-

ical solution of the wave-equations. Compu-

tational simulations of the electrical behavior

of the travelling-wave photodetector are car-

ried out using an equivalent circuit model.

IntroductionThe electrical behavior of the travelling-wave

photodetector can be described using the equiv-

alent circuit model of Fig. 1 [1]. The travelling-

wave photodetector is fabricated as an electri-

cal millimeter waveguide. Therefore all elements

are per unit length.


∂

u3

i B

I z dzp’ ( )

Y dz3’

u1

u2Y dz2’

I z dzp + ’ ( )i A

Y dz1’

i A

W dz’i

u u uz

+ ∂∂

i iz

dz+ ∂

Fig. 2: Summarized equivalent circuit model of the travelling-wave photodetector. All elements are per

unit length.

The impedances Rm and Rhl describe the

longitudinal ohmic losses in the metalization and

the semiconductor, respectively. The inductance

of the electrical waveguide is taken into account

with the inductance L. The conductance Grlzand the capacity Crlz consider the Schottky-con-

tact depletion layer. The two elements Cb and

Gb describe the behavior of the bulk material.

An additional capacity CL is introduced for the

electric field in air above the photodetector. The

impressed current source IPh describes the opto-

electric conversion in the absorbing quantum

well layer.

Numerical solutionThe wave-equations of the travelling-wave

photodetector are derived by using a summa-

rized equivalent circuit model, shown in Fig. 2,

which based on the model above. The equation

of the complex voltage amplitude along the

transmission line can be shown as

∂∂

2

21 2 1 3 2 3

1 2

u

zu W

Y Y Y Y Y Y

Y Y= ⋅ ⋅ ⋅ + ⋅ + ⋅

+’

’ ’ ’ ’ ’ ’

’ ’

2

1 2

I zW Y

Y Yp+ ⋅ ⋅

+’

’ ’

’ ’( ) (1)

where W ist the impedance and Y1, Y2 and Y3the admittances of the transmission line. Ip(z)

is given by

I zq R P

hpopt opt opt’ ( )

( ) =

⋅ ⋅ − ⋅ ⋅⋅

η αν

1

j zopt opt( )e⋅ − + ⋅α β

(2)

where hopt is the internal quantum efficiency, aopt

is the optical absorption coefficient, hn is the

photon energy, Popt is the incident light power,

R is the reflection of interface device/air and bopt

is the optical phase coefficient.

3 RESEARCH18

0 0,2 0,4 0,6 0,8 1z (mm)

0

5

10

15

20

25

30

35

40

Vo

ltage

(m

V)

Fig. 3: Voltage distribution along z direction for a simplified

(interrupted line) and the fully equivalent circuit model.

Using the finite differences method [2], the

numerical solution of equation (1) is given by

u

u u h I zW Y

Y Y

h WY Y Y Y Y Y

Y Y

k

k k p k

=+ − ⋅ ⋅ ⋅

+

+ ⋅ ⋅ ⋅ + ⋅ + ⋅+

+ −1 12 2

1 2

2 1 2 1 3 2 3

1 2

2

’’ ’

’ ’

’’ ’ ’ ’ ’ ’

’ ’

( )

(3)

with

hb a

N = −

(4)

where a is the beginning and b the end of the

transmission line. N is the number of discrete

points, where the voltage is calculated

(0 £ k £ N).

The loads at the two ends of transmission line

are defined as Z1and Z2, the characteristic re-

sistance of the transmission line is Z. According

to microwave theory there exist reflections r1 and

r2 at the two ends z = 0 and z = l of the device.

The boundary condition for this problem is de-

termined by the current voltage relation at the

specific load resistance.

u z u( )= =0 0

[ ]Z

W

r

r hu u’= ⋅

+−

⋅ ⋅ − −1

11

21

11 1

(5)

u z l uN( )= =

[ ]Z

W

r

r hu uN N’= − ⋅

+−

⋅ ⋅ −+ − 1

11

22

21 1

(6)

The solution of equation (3) is computation-

ally calculated.

SimulationFor the simulation of photoelectrical micro-

wave generation a frequency of 40GHz is tak-

en. The wavelength of the optical sources is

1.3µm and the light power of the two beams are

both 1mW. In the following calcu-

lation the quantum efficiency hopt

of photoelectrical conversion is

assumed to be 1 and the incident

light energy is fully and uniformly

coupled to the active layer, so the

optical reflection R is 0.

Fig. 3 shows the voltage dis-

tribution along z direction. The

reflection factor at z = 0mm is 1

and at z = 1mm the reflection is

0.

The dashed line shows the re-

sult by using a simplified equiva-

lent circuit model, in which Cb, Gband CL are neglected. An exist-

ing simulation program, which

calculates the solution analytical-

ly, is based on this model. The


0 0,1 0,2 0,3 0,4 0,5z (mm)

0

10

20

30

40

50

60

Vo

ltag

e (m

V)

Fig. 4: Voltage disribution along z direction for Z2 = 50W

(dashed line) and Z2 = Z (solid line) for a device length of

comparison with the numerical simulation, which

is based on the complete equivalent circuit mod-

el, shows, that the efficiency is reduced by 10%.

One possibility to optimize the device rela-

tive to the output-voltage is to reduce the length

of the transmission line. In the following simula-

tion the length of the device is set to 0.5mm.

Fig. 4 shows the results by using Z 2 = Z (solid

line) and Z2 = 50W (dashed line). In case of ad-

aptation the efficiency is raised by 44% relative

to a device length of 1mm. In case of mismatch-

ing there is a high voltage at the end of the de-

vice, but the efficiency has not raised because

of higher load at the device´s end.

Another possibility of optimization is to reduce

the value of Rm. Simulations have proved that

the efficiency is raised by 20%, if Rm is set at

0W (device length is 1mm).

ConclusionThis report presents the nu-

merical solution of the photoelec-

trical microwave generation of a

coplanar travelling-wave photo-

detector. The wave equations are

solved by the finite differences

method. The results of the simu-

lations have shown, that the effi-

ciency is reduced by 10% relative

to using a simplified equivalent

circuit model and that a further

optimization is possible.

References[1] D. Jäger, R. Kremer, Trav-

elling-wave optoelectronic devices

for microwave applications, Proc.

IEEE, MTT-S and LEOS Topical

Meeting on Optical Microwave Interactions, pp.

11-14, 1994, France

[2] D. Marsal, Finite Differenzen und Elemente:

numerische Lösung von Variationsproblemen

und partiellen Differentialgleichungen, Springer-

Verlag, Berlin/Heidelberg, 1989

3.1.4 Determination of RF-equivalentcircuit elements of travelling-wavephotodetectors using network analy-sis

O. BERGER AND M. ALLES

n this report, the determination of the

elements of the equivalent circuit model

of coplanar waveguides is described. Since

the longitudinal and the transverse complex

3 RESEARCH20

I

Fig. 1: Equivalent circuit model of the travelling-wave photode-

tector.

impedance of the equivalent circuit model

can be calculated from the characteristic im-

pedance and the propagation coefficient

measured with a network analyzer, it is pos-

sible to compute the equivalent circuit

straight forward. This method has been used

to determine the equivalent circuit model of

travelling-wave photodetectors. Results and

comparison to theory and other measure-

ments are shown.

Introduction

Recently, 60GHz-travelling-wave photodetec-

tors are under development in the Fachgebiet

Optoelektronik. For characterization and further

optimization of the device, network analyzer RF-

measurements are analyzed in a new way in

order to determine the high-frequency equiva-

lent circuit elements of the device. The imple-

mented method determines the equivalent cir-

cuit model directly from network analyser mea-

surements.

Equivalent Circuit ModelThe coplanar waveguide structure of the trav-

elling-wave photodetector can be described us-

ing the distributed equivalent circuit model

shown in Fig.1 [1]. Note that all elements are

per unit length. The impedances RM and R

describe the longitudinal ohmic losses in the

metalization and the semiconductor, respective-

ly. The inductance of the electrical waveguide

is considered by the inductance L. The deple-

tion layer of the Schottky-contact is taken into

account with the conductance G and the ca-

pacitance C while GB and CB characterize the

bulk material of the semiconductor. The addi-

tional capacitance CL is associated with the

electrical field in air above the device. Finally,

the impressed current source IPh is introduced

to describe the optoelectronic conversion with-

in the absorbing layer.

To determine the equivalent

circuit elements it is necessary to

make some simplifications. Usu-

ally, network analyser measure-

ments take place without optical

illumination of the device, there-

fore IPh can be neglected. The

two capacitances CL and CBhave much less influence on the

behavior of the device in compar-

ison to the Schottky-capacitance

C and can be neglected.

One can conclude, that in the

longitudinal and in the transverse

part of the equivalent circuit three

elements are to be considered:

two real impedances and one

imaginary one. The longitudinal


theory low-frequency high-frequencymeasurements measurements

RmmM’Ω

6.58 6.21 13.8

Rmm

’Ω

707 - 887

GS

mm’

6.7×10-14 750×10-12 1.3×10-3

GS

mmB’

874 1.1 2.54

LnHmm

’

0.466 - 0.62

CpF

mm’

1.39 1.17 0.91

Tab. 1: Values for the elements of the equivalent circuit model

of a travelling-wave photodetector.

and the transverse part of the

equivalent circuit can be de-

scribed separately with

WR R j L

R R j LM

M’

’( ’ ’)

’ ’ ’= +

+ +ωω

and

.Y

G G j C

G G j CB

B’

’ ( ’ ’)’ ’ ’

= ++ +

ωω

Using separate Smith charts

for the longitudinal and the trans-

versal part of the equivalent cir-

cuit to display the frequency de-

pendence of W and Y,

semicircles with two intersections

with the real axis for ω = 0 and

ω → ∞ can be determined. The

intersections with the real axis are

ZR R

R RM

M’( )

’ ’’ ’

ω = =+

0 ,

Z R’( ) ’ω → ∞ =

for the longitudinal part and

YG G

G GB

B’( )

’ ’’ ’

ω = =+

0 ,

Y G B’( ) ’ω → ∞ =

for the transverse part.

Determination of the equivalent circuitelements

The network analyser measures S-parame-

ters from a device under test (DUT). The char-

acteristic impedance Z and the propagation co-

efficient g are determined from this data.

The new method calculates the impedance

of the longitudinal part and the admittance of

the transverse part using the relations W = g Zand Y = g / Z. The frequency dependence of

both parts is displayed in separate Smith charts.

An statistic-based algorithm fits a semicircle to

the measured data. The intersections with the

real axis are used to determine the real elements

of the equivalent circuit model. With these re-

sults, it is possible to calculate the imaginary

elements L and C for each frequency. This

method has been implemented to an easy-to-

handle windows-program with graphic features.

The calculation results are displayed instantly

on-screen.

ResultsThe equivalent circuit elements calculated

using this method have been compared with

results of low-frequency measurements and the-

oretical determined values [2], Tab. 1. As can

3 RESEARCH22

0 10 20 30 40

frequency (GHz)

40

30

20

10

0

−10

−20

−30

reZ

imZ

Fig. 2: Characteristic impedance of a travelling-wave

photodetector.

4

3

2

1

0

4

3

2

1

00 10 20 30 40

frequency (GHz)

Fig. 3: Phase coefficient and attenuation coefficient of a

travelling-wave photodetector.

be seen from this table, the measure-

ments are in good agreement with the-

oretically determined values. Only the

two admittances show different values.

In case of G the network analyzer mea-

surement leads to a higher value. An

explanation is, that the accuracy of the

network analyzer makes it impossible

to measure these low admittances.

The theoretical value of the bulk ad-

mittance GR is much larger than the

measured values. A reason is, that the

metal-semiconductor resistance is ne-

glected in the derivation of GB.

Taking measurements at various

bias voltages one can see that only the

elements C and G regarding the de-

pletion layer show major bias depen-

dence while the other ones keep con-

stant over the bias voltage.

Fig.2 shows the characteristic im-

pedance, Fig. 3 the phase and attenu-

ation coefficient calculated from the

measured data. The real part of the

characteristic impedance rises at fre-

quencies above 35GHz indicating that

probably the major part of the electri-

cal quasi-TEM wave on the structure

changes from TEM to TM.

The smithcharts with the measured

data and the semicircle calculated as

an approximation for the longitudinal

part and the transversal part of the

equivalent circuit are shown in Fig. 4.

As is visible, the frequency dependence

of the measured data leads to semicir-

cles, indicating that the device under

test can be described with the simpli-

fied equivalent circuit model described

above.


(a)frequency

(b)

frequency

Fig. 4: Smithcharts for the longitudinal part (a) and for the transversal part (b).

frequency (GHz)

0 10 20 30 40

1.0

0.8

0.6

0.4

0.2

0.0

1.0

0.8

0.6

0.4

0.2

0.0

Fig. 5: Inductance and capacitance versus frequency.

The frequency dependence of the inductance

L and the capacitance C, shown in Fig. 5, can

be determined directly with the program. Due to

a suggested change of the quasi-TEM-Mode to

a TM-mode, the inductance rises and the ca-

pacitance decreases with higher frequencies.

ConclusionsA method to determine the RF-equivalent el-

ements directly from network analyser measure-

ments has been developed. This method ap-

proximates the frequency dependence of the

measurement data in a Smith chart graphically

without any further knowledge about the values

to be determined. The measured results fit well

with both theoretically determined and low-fre-

quency measured values. It is also possible to

examine the frequency dependence of the in-

ductance and the capacitance.

AcknowledgementThe author would like to thank U. Auer (Fach-

gebiet Halbleitertechnik/-technologie) for grow-

ing the epilayer and for fabrication of the travel-

ling-wave photodetector.

References[1] M. Alles, T. Braasch, D. Jäger,

High-speed coplanar Schottky trav-

elling-wave photodetectors, Int. Conf.

on Integrated Photonics Research,

Proc. pp. 380-383, Boston, USA, 1996

[2] O. Berger, Bestimmung der HF-

Ersatzschaltbildelemente von Photo-

detektoren mit Hilfe der

Netzwerkanalyse, Graduate thesis,

Fachgebiet Optoelektronik, Gerhard-

Mercator-Universität Duisburg, 1997

3 RESEARCH24

3.1.5 Polarization insensitivewaveguide modulators on InP

T. ALDER AND R. HEINZELMANN

lectroabsorption modulators using

strained multiple quantum well (MQW)

structure have been designed, fabricated and

characterized. Utilizing the Quantum Con-

fined Stark Effect (QCSE) due to high elec-

tric field underneath a Schottky-electrode,

the absorption coefficient of the optical

waveguide can be changed. The use of

strained quantum wells enables an operation

of the device, with almost no sensitivity to

different polarisation. With this device an on/

off-ratio of 18.5dB has been achieved.

IntroductionIn general, the absorption change in a MQW

structure is strongly polarization dependent [1].

From the viewpoint of system applications, a

polarization insensitive or at least polarization

independent modulator is desirable. Appropri-

ate structures can be designed using quantum

wells with tensile strain [2-3]. In this paper elec-

troabsorption waveguide modulators using a

strained InGaAs/InAlAs MQW structure in the

electrooptical active region will be presented.

Device structure and principle of operationA schematic diagram of the modulator struc-

ture is shown in Fig. 1. The modulators investi-

gated utilize a nin-structure containing Si-doped

InAlAs top and bottom cladding layers with thick-

ness of 570nm and 1120nm, respectively.


Fig. 1: Schematic diagram and cross section of the modulator.

E

0V -2V -4V -6V -8V -10V

position position position position position position

Fig. 2: Nearfield-pattern and lateral profile of

the optical waveguide mode at different

reverse biases.

-10 -8 -6 -4 -2voltage [V]

0

20

40

60

80

100

tran

smis

sion

[%]

= 1,2mm

w = 14µm

Fig. 3: Transmission as a function of different

reverse biases.

The doping concentration of the bottom clad-

ding layer is ND = 1×1017cm-3, whereas that of

the top cladding layer is ND = 1×1016cm-3. The

non intentionally doped guide consists of 19 ×

6nm thick InGaAs MQWs separated by

19 × 7.7nm thick InAlAs barriers. The structure

was grown using the MBE machine of the De-

partment of Optoelectronics. To examine the

dependence of the device behaviour on contact

geometry, a number of devices were fabricated

with chrome-gold-Schottky-electrodes of differ-

ent width, ranging from 8µm to 16µm. The Schot-

tky-electrodes were manufactured by thermal

evaporation of chrome and gold in ultra high

vacuum. The waveguide structure was formed

using wet chemical etching after evaporation of

the Schottky contacts. The second contact,

shown in Fig. 1 is carried out as Ohmic contact

and was manufactured by thermal evaporation

of germanium, nickel and gold in ultra high vac-

uum.

As a reverse bias is applied to the Schottky-

electrode, there will be a high electric field un-

derneath the Schottky-contact within the deple-

tion region. This increases the absorption

coefficient of the guide due to the quantum con-

fined Stark effect (QCSE). In this way, the opti-

cal output power can be controlled electrically.

Experimental resultsFig. 2 shows the nearfield-pattern and the

lateral profile of the optical waveguide mode at

different reverse biases. From this figure, it can

be seen, that the output becomes weaker as

the applied reverse bias is increased. This be-

havior is due to the increased absorption coeffi-

cient in the optically guiding region.

As the previous result shows, it is possible to

control the optical output power by a reverse

bias. In the following, systematic results on trans-

mission changes will be presented. In Fig. 3 the

transmission is plotted as a function of different

reverse biases. From this figure, it can be seen,

that within the range from -4.3V to -7.4V the

transmission changes almost linear with the

applied bias. The ratio between maximum and

minimum transmission is 18.5dB. Furthermore

it can be seen, that the change in transmission

from 0V to about -4V is very low. This indicates

that quantum wells were grown smaller than they

were designed.

Fig. 4 shows the transmission as a function

of reverse bias for TE- and TM-polarization. It is

evident from the figure, that for TE- and TM-po-

3 RESEARCH26

7 9 11 13 15 17 contact width [µm]

0

4

8

12

16

20

max

imum

tra

nsm

issi

on c

hang

e [d

B]

TE - polarisation

TM - polarisation

Fig. 5: Maximum transmission change as a function of different

reverse biases for TE- and TM-polarization.

-10 -8 -6 -4 -2voltage [V]

0

0,2

0,4

0,6

0,8

1

tran

smis

sion

[a.

u.]

18,5 dB

17,2 dB

TE - polarisation

TM - polarisation

Fig. 4: Transmission as a function of different reverse biases for

TE- and TM-polarisation.

larization the change in transmis-

sion is almost equal. While for TE-

polarization an on/off-ratio of

18.5dB could be measured a

slightly smaller value of 17.2dB

appeared for TM-polarized light.

Considering only the linear

range (the voltage range be-

tween -4.3V and -7.4V bias volt-

age) of the transmission charac-

teristic, for TE- as well as

TM-polarization an on/off-ratio of

8.2dB is measured.

To examine the dependence

on device dimensions modulators

with different Schottky-electrode

widths from 8µm to 16µm were

investigated. The results are

shown in Fig. 5. As can be seen,

there is no recognizable influence from contact

width on the maximum transmission change, nei-

ther for TE-polarization nor for TM-polarization.

This is of major importance, as with a change in

the contact width the propagation

properties of the electrical

waveguide can be fit to those of

the optical waveguide, for high-

speed operation the travelling-

wave concept [4] can be applied.

ConclusionsElectroabsorption waveguide

modulators based on strained In-

GaAs/InAlAs-MQW have been

designed, fabricated and charac-

terized. A maximum on/off-ratio of

18.7dB has been achieved. It

could be shown, that the polari-

sation influence on the transmis-

sion behaviour was small, due to

the influence of the strain in the

quantum well region. Additionally


no influence of the contact dimensions on the

transmission was observed.

References[1] T. Aizawa, K. G. Ravikumar, R. Yamauchi,

Polarisation Independent Refractive Index

Change In InGaAs/InGaAsP Tensile Strained

Quantum Well , Electronics Letters, Vol. 29, No.

1, pp. 21 - 22, January 1993

[2] H. W. Wan, T. C. Chong, S. J. Chua, Consider-

ations For Polarisation Insensitive Optical Switch-

ing and Modulation Using Strained InGaAs/

InAlAs Quantum Well Structure, IEEE

Photonics. Techn. Lett., Vol. 3, No. 8, pp. 730 -

732, August 1991

[3] J. Shimizu, T. Hiroshima, A. Ajisawa, M.

Sugimoto, Y. Ohta, Measurement of the

polarisation dependence of field induced refrac-

tive index change in GaAs/AlAs multiple quan-

tum well structures, Appl. Phys. Lett., Vol. 53,

No. 2, pp. 86 - 88, 1988

[4] D. Jäger, R. Kremer, and A. Stöhr, Travelling-

wave optoelectronic devices for microwave ap-

plications, IEEE MTT-S 1995 International Mi-

crowave Symposium, Vol. 1, pp. 163-166, 1995

(invited paper)

3 RESEARCH28

Fig. 1: Signal and energy transmission into the eye

3.2 Optical Interconnects andProcessors

3.2.1 Neurotechnology: Retina Im-plant

M. GROSS AND R. BUSS

he Department of Optoelectronics is

a member of a consortium of 14 Ger-

man expert groups, working on the project

EPI-RET: Retina Implant. This interdiscipli-

nary project, funded by the Federal Ministry

for Education, Science, Research and Tech-

nology (BMBF) in Germany, is developing a

retina implant.

This device is a neural prosthesis, designed

for patients blinded by a disease where the

outer retinal layer degenerates (retinitis pig-

mentosa or macula degeneration). It consists

of three parts: a so-called retina encoder (RE)

outside the eye, simulating the function of

the retina, the retina stimulator (RS), a mi-

crochip placed on the retina with electrodes

stimulating the ganglion cells in the outer

retina layer, and a wireless signal and ener-

gy transfer from RE to RS.

The task of the Department of Optoelectron-

ics in this project is the development of a

device for optoelectronic signal and energy

transfer into the eye. To achieve this, a pro-

totype consisting of a laserdiode as trans-

mitter and a receiver consisting of a

monolithically integrated photovoltaic cell

array and a photodiode, together with driv-

ing and receiving electronics, was manufac-

tured.

IntroductionThe signal and energy transmission line de-

scribed here is part of a technical system func-

tioning as a vision aid for blind people who have

lost their vision due to retinal degenerations,

especially retinitis pigmentosa [1]. An often ap-

pearing kind of blindness is the partially degen-

eration of the retina, e.g. the disease retinitis

pigmentosa, which is leading to blindness

through following steps: The typical begin is the

loss of the rod photoreceptors, causing night

blindness. Next the cone photoreceptors are

3.2 Optical Interconnects and Processors 29

T

dying off, beginning at the outer perimeter of vi-

sion. This leads to a tunnel vision and finally to

total blindness, when the cones in the fovea are

lost.

However, while the photoreceptors are dying

off, the nerve cells in the retina and subsequent

parts of the central visual system are remaining

mostly intact [1]. This leads to the possibility of

developing a visual prosthesis with the ability to

replace main parts of the retina that gives sight

back to the visually impaired [2].

System description The whole system sketched in Fig. 1 con-

sists of three main parts:

> a retina encoder (RE), consisting of a CMOS

camera and an artificial neural network (en-

coder) for image data processing,

> a retina stimulator (RS), a flexible chip, epiret-

inal affixed, with µ-electrodes on the back,

> a wireless signal and energy transmission line

from the retina encoder to the retina stimula-

tor.

The system works as follows: First the high

dynamic range CMOS camera generates a pic-

ture. This dataset is then reduced by an artifi-

cial neural network and transformed into digital-

ly coded pulse trains (nerve signals), to which

the ganglion cells can react. This corresponds

to the data reduction from 120 million photore-

ceptors to 1 million ganglion cells by the human

retina. The dataset is then optically transmitted

at a rate of 1 Mbit/s to a microcontact foil on the

retina (receiver), where eye movements of up

to +/- 15°, measured from looking straight ahead

have to be compensated. Together with this in-

formation transfer an optical bias is transmitted,

supplying the driving circuit with 5 mW electri-

cal power. The retina stimulator is a soft micro-

contact foil which is implanted adjacent to the

ganglion cell layer on the outer retinal limit. The

µ-electrodes are stimulating the ganglion cells

of the retina, thus transmitting the signals via

the optic nerve to the visual cortex in the brain.

ResultsThe signal and energy transmission has been

realized in a first prototype using a laser diode

as transmitter and a photovoltaic cell array to-

gether with a photodiode as receivers for ener-

gy and signals, respectively. In the first step we

designed the parts for the optical transmission,

considering the boundary conditions given by

the human eye and the technical demands of

the whole device. For surgical reasons optical

fibres cannot be used to connect transmitter and

receiver directly. Therefore, a light source (e.g.

a pigtailed laser diode) has to be fixed in front of

the eye, transmitting the signal and energy onto

the retina, using free space optics. A micro-lens

system was developed mapping the light homo-

geneously on the retina in a spot with a diame-

ter of about 5 mm. This assures that the receiv-

er is illuminated for eye movements of up to +/

- 15°. The material used for the receiver is GaAs,

mainly to achieve high conversion efficiencies

with the photovoltaic cell array [2]. This has sev-

eral advantages:

1. The fibre has an gaussian beam profile, while

laser diodes have strong astigmatism that has

to be corrected with a microoptic in front of

the eye.

2. The heat that the laser diode produces is led

away very easily.

3. The high frequency modulation of the laser

diode for the signal transmission is made far

away of the eye avoiding problems with elec-

tromagnetic compliance.

Latest results are shown in Fig. 2 and Fig.3:

Fig. 2 schematically depicts the system design.

3 RESEARCH30

Fig. 2: System design

In Fig. 3(a) the I-V characteristics of a single pho-

tovoltaic cell and of an array with 5 cells con-

nected in series are plotted. At a wavelength of

l = 800 nm this array delivers up to 5 mW elec-

trical power with a conversion efficiency of about

23%, which turns out to be a great improvement

as compared with the results published in [3]. It

should be noted, however, that this efficiency

was obtained without any antireflection coating.

Experiments have shown that the efficiency can

be increased up to almost 31% by encapsulat-

ing the cell array with a biocompatible antire-

flection coating consisting of SiO2/Si3N4 multi-

layers. Moreover, Fig. 3(b) shows

measurements of the signal transmission: The

signals at the output of the encoder, curve (1),

are transmitted optically into the eye at a rate of

1 Mbit/s. The output of the receiver is plotted in

curve (2) of Fig. 3(b) together with the recovered

clock, curve (3), in Fig. 3(b).

ConclusionIn this report the progress in the work for an

optoelectronic signal- and energy transmission

line for use in a visual prosthesis is presented.

A concept is developed and a prototype is de-

scribed. This prototype currently has the capa-

bility of delivering 5 mW electrical power together

with digitally coded signals at a rate of 1 Mbit/s

simultaneously, thus meeting the current sys-

313.2 Optical Interconnects and Processors

Fig. 3: (a) I-V characteristic of photovoltaic cells (PVCs), (b) digitally coded signals before (1) and

after (2) transmission, and (3) recovered clock signal.

tem requirements. However, the optical link de-

scribed here is capable of transmission rates up

to 1 Gbit/s, suitable for optically powered high-

speed data links.

AcknowledgementThe authors would like to thank the Federal

Ministry for Education, Science, Research and

Technology for financial support and all mem-

bers of the EPI-RET team for fruitful discussions.

References[1] R. Eckmiller Retina implants with adaptive retina

encoders, Proc. of the 1996 RESNA Research

Symp., Salt Lake City, pp. 21-24, 1996

[2] M. Groß, T. Alder, R. Buß, R. Heinzelmann, M.

Meininger, and D. Jäger, Micro Photovoltaic Cell

Array for Energy Transmission into the Human

Eye, Proc. of the 14th European Photovoltaic So-

lar Energy Conference, Barcelona, Spain, vol.

1, pp. 1165-67, 1997

[3] J. Rizzo, J. Wyatt, Silicon retinal implant to aid

patients suffering from certain forms of blind-

ness, Proc. of the 1996 RESNA Research

Symp., Salt Lake City, pp. 1-3, 1996

3.2.2 Analysis of the optical energyand signal transfer module for anartificial vision prosthesis

T. BAUMEISTER, M. GROSS, AND R. BUß

ithin the scope of the Retina Im-

plant Project supported by the Ger-

man government the possibility of a wireless

transfer of signal and energy into the eye of

human patients was analyzed.

IntroductionThe Retina Implant project was founded in

1995 as part of a young and interdisciplinary

area of research: Neurotechnology.

The goal of this ten year project is the devel-

opment of both an artificial eye implant (retina

stimulator), stimulating the ganglion cells of the

human retina from patients, who lost their eye-

sight due retinitis pigmentosa or macula degen-

eration and an encoder transforming the signals

coming from a video system like human retina

does.

One of the coming tasks is to develop a sys-

tem transporting the electrical power for this

implant and the signals from the output of the

encoder into the eye. This system analysis

shows the technical preferences for this trans-

port by IR-rays.

Criteria catalogueThe base of each scientific analysis is a cri-

teria catalogue being a decisive help for the

evaluation of possible alternatives. The follow-

ing criteria were found:

1. The fundamental criteria are the dimensions

of the implant. Due to surgical reasons the

maximum length is limited to 1.5 mm.

2. The Efficiency of the power transmission is a

criterion of great importance for any implant-

ed system, because of the absence of possi-

bility to cool any part of system inside the eye.

3. The reliability is fundamental too, because it

is nearly impossible to repair any failure and

the exchange of the whole system is more

dangerous for the patient as the first time im-

plantation.

4. The biocompatibility is one more very impor-

tant criterion for the long time function of any

3 RESEARCH32

W

implant. This may be given by a biocompati-

ble coating of any material, but there is the

risk of damage during the implantation and

fixation of the innerocular part. Further, we

should be aware of the influence of overgrow-

ing the receiving part of the signal and ener-

gy transport system.

5. The receiver can´t be placed at all places on

the retina, because the fixation of it may dam-

age axons of stimulated ganglion cells.

6. The technical availability and the need of

development of parts of the system are im-

portant due to the cost and the time to mar-

ket of the system.

7. The possibility of extension, especially the

number of stimulating electrodes, is an im-

portant criterion for the future.

8. The acceptance of the whole system by the

patient and his milieu is quiet important too.

System analysisFirst the need of bandwidth and power for the

stimulation of a given amount of stimulating elec-

trodes were analyzed. The first version of the

retina stimulator will consist of an

array of twenty stimulating elec-

trodes.

The bandwidth needed for the

stimulation with twenty electrodes

is nearly 100kbit/s, for 400 elec-

trodes we found a bandwidth of

approximately 25Mbit/s.

This calculation of the band-

width includes the scheme shown

in Fig. 1, the number of bits need-

ed to address the electrodes, the

number of bits needed for encod-

ing the stimulating pulseform, the

rate of neuro impulses, and fac-

tor needed for encoding the signal by wireless

transmisson.

The power needed using a single amplitude

modulated IR-laser for simultaneous transport

of energy and signals is 190mW of optical pow-

er in a worst case analysis with twenty electrodes

stimulating in one time frame. The worst case is

defined here by a constant electrical stimula-

tion power of 750µW at the electrodes.

Next, the possibilities of signal encoding were

analyzed. The whole signal of amplitude modu-

lated laser beam includes an AC signal, the en-

coded signal for stimulation and addressing the

electrodes, and a DC offset for the energy trans-

fer.

The CMI-code (see Fig.2) also known as 2-

AMI-1-code or modified FSK-code is the best

compromise between the technical expense of

signal and clock recovering and the need of a

DC free signal in this application. This coding is

mentioned with a factor of two in analysis of

bandwidth [1]. Considering this we found the

best way to transfer the energy and signals is to


Fig. 1: Stimulation scheme

use a single amplitude modulated IR-laser di-

ode.

Analysis of the opticsTo avoid a complicated, heavy and cost in-

tensive eyetracking system focussing the laser-

beam exactly on the position of the energy and

signal receiving photodiode, we decided to illu-

minate an area on the retina great enough to

compensate eyemovements about 10° in each

direction. The position of this area is vertical

ahead the macula. At this position the beam is

least influenced by the eye lid and the greatest

eye movement possible. The best way to lower

the reflection of the beam at the iris and thereby

to expand the possible angle of movement is to

focus the beam in the hole of the iris. For a nearly

uniform illumination of area, needed for an uni-

form supply of power of the stimulation electron-

ic, we have developed a ring-shaped focus of

the beam. Thus, we solved the problem how to

3 RESEARCH34

Fig. 2: Calculation of the power needed

Fig. 3: Example of a DC-free encoded signal

added with an offset for transmision of power

get a uniform illumination on the screen of a good

imaging optical system i.e. human eye. To solve

this problem we simulated the whole optical path

with the ray tracing program ZEMAX SE for hu-

man eye and additional for the rabbit eye, see

Fig. 5. For the simulation of the human eye we

used a slightly modified model of the eye from

Helmholtz. We completed this model with the

axes of movement [2]. For the eye of the rabbit

we used a similar model [3].

The optics consists of a two lens beam ex-

pander and one complex collecting lens to get

the above mentioned ring shaped focus (see

Fig.5). The optics is not modular, i.e. it is not

possible to use parts of the optics designed for

the human eye in the optical system for the rab-

bit eye.

The optical source can be the end of an opti-

cal fiber or a laser diode with a micro lens that

corrects the astigmatism of the laser. The lens-

es used in our model are all in the range of to-

days industrial optics including one standard

micro lens.

ConclusionHere is shown, that, principally, it is possible

to transfer enough power and a signal with suf-

ficient bandwidth into the eye by using an IR

light beam. Further it is shown, that it is prefera-

ble to use a single amplitude modulated laser

diode according to the above mentioned cata-

logue of criteria. This paper shows that a trans-

port of power and signal from the signal pro-

cessing unit outside the eye to the implanted

microelectronics by optical means can be real-

ized without the need of an eye tracking sys-

tem.

3.2.3 Development of an optical sig-nal and energy transmission system

R. HEDTKE AND M. GROSS

n optical transmission system to pow-

er and provide a retinal implant with

energy and digitally coded information is de-

veloped.

IntroductionThe EPI-RET: Retina

Implant project is part of the

Neurotechnology Program

of the Federal Ministry for

Education, Science, Re-

search and Techno-

logy (BMBF). The implant

system is evolved in co-op-

eration with several interdis-

ciplinary project partners as

a vision aid for people who


Fig. 4: Example of CMI encoding

Rabbit eye Human eye

Fig. 5: Optical analysis of the rabbit eye and the human eye

A

are suffering from retinal degenerative defects

like retinitis pigmentosa and macula degenera-

tion. With the help of this optical transmission

system the retinal implant is provided with the

needed information and energy.

PrincipleFor the transmission to the ret-

inal implant the information is

modulated onto the laser light,

which transmits the energy. The

kind of modulation that is used is

shown in Fig. 1. A continuos en-

ergy transmission into the implant

is guaranteed by modulating the

driving current around a mean

operating point (OP). The driving

current and the optical output-

power for the laser diode is

sketched for an stimulation with

a rectangular-signal.

TransmitterThe driving current, as shown

in Fig.2, is regulated by a current

control unit to secure a constant

optical output power of the laser diode. This con-

trol unit compares the preset current value with

the actual and provides the control signal for the

driving unit. To generate the actual current val-

ue a photodiode integrated into the laser diode

[1] (alternatively: an external shunt-resistor) is

used. With the help of a signal conversion the

measured signal is processed for the following

current control unit. The modulation of the driv-

ing current occurs in the previously described

way.

ReceiverThe modulated laser light hits the photodiode

and the photovoltaic cell array (see Fig.3). The

photodiode generates the detector signal which

is transformed to TTL-level by a signal process-

ing unit. A clock recovery unit is used to gener-

ate the clock signal. The other part of the laser

3 RESEARCH36

DRIVING CURRENT (I )I th

TIME

TIM

EOP

TIC

AL

OU

TP

UT-

PO

WE

R (

PO

PT )

I

PO

PT

23

Fig. 1: Principle of the modulation of the

laser diode

LASERDIODE

CURRENT ADJUSTMENTCURRENTCONTROL

DRIVINGUNIT

OPTICAL OUTPUT-POWER

PHOTODIODE

AND

SIGNALCONVERSION

SIGNAL MODU-LATOR

DR

IVIN

G C

UR

RE

NT

+

ALTERNATIVE GENERATION OFACTUAL CURRENT VALUE

RESISTORAND

SIGNALCONVERSION

MO

DU

LAT

ION

CU

RR

EN

T

ACTUAL CURRENT VALUE

Fig. 2: Block diagram of the transmitter

light is converted into electrical energy by the

photovoltaic cell array [2].

Measured resultsFig. 4 shows the driving current of the laser-

diode for a stimulation with a 1 MHz rectangular

wave. The used mean operating point is 400

mA [3] or rather 190 mW optical power. To code

the signal the Manchester code [4] is used. The

main advantage of this coding are the well-bal-

anced relation of logical high an low states (im-

portant for a constant energy transmission) and

the easy recovery of the clock signal. The trans-

mitted and the received Manchester coded sig-

nal and the recovered clock are shown in Fig. 5. ConclusionAt the Fachgebiet Optoelektronik (Gerhard-

Mercator-Universität Duisburg) an optical signal

and energy transmission system was developed

to provide a retinal implant with digitally coded

signals. To realize this aim a transmission and

a receiving unit were designed.

References[1] SDL, Laser Diode Operators Manual & Techni-

cal Notes, SDL Inc., San Jose/USA, 1994

[2] M. Meininger, Entwicklung photovoltaischer

Zellen zur Energieversorgung einer künstlichen

Sehprothese (Retina Implantat) Diploma thesis,


1 2 3 4 50

200

400

600

800

t/µs0

I/mA

Fig. 4: Modulated driving current for the

laserdiode

PHOTO-VOLTAIC

CELLARRAY

SIGNALPHOTODIODE

SIGNAL PRO-

CESSING

CLOCKRE-

COVERVY

CLOCK

MODULATED LASER LIGHTDETECTOR

SIGNAL

ENERGY

Fig. 3: Block diagram of the receiver

0 5 10 15 20

CLOCK

RECEIVED SIGNAL

TRANSMITTED SIGNAL

t/µs

Fig. 5: In- and output signals of the transmis-

sion system

Fachgebiet Optoelektronik, Universität Duisburg,

1997

[3] T. Baumeister, Systemanalyse des optischen

Energie- und Signalübertragungsmoduls für eine

künstliche Sehprothese (Retina Implantat) Di-

ploma thesis, Fachgebiet Optoelektronik,

Universität Duisburg, 1996

[4] R. Mäusl, Digitale Modulationsverfahren,

Hüthing, Heidelberg, 1991

3.2.4 Infrared data link for rotatingdisplay-systems

U. WEIMANN AND R. BUß

he magicball is a recently designed

rotating LED display system for texts

and graphics. In this report two methods of

wireless programming of the magicball by

using either an infrared remote control or a

notebook-PC are presented. In the first in-

stance, a polymethylmethacrylate (PMMA)-

body is designed and a new software is

developed allowing the programming via an

IR-remote control unit. An infrared interface

(dongle) for notebook-PCs without an built-

in infrared interface and a special data trans-

mission software protocol is described in the

second part of the report.

IntroductionThe magicball display-system is used as an

eye-catcher in stores or at exhibitions amongst

others. Its design and operation principles are

shown in Fig. 1. The 16 LEDs are fixed at the

end of the rotating arm. The microcontroller cre-

ates moving titles on the surface of the mag-

icball by switching the LEDs on and off individ-

ually. The microcontroller and the LEDs on the

rotating carrier and arm are powered by a gen-

erator situated in the socket of the magicball.

The data transmission was formerly achieved

by a serial PC cable and rubbing contacts. By

designing a wireless infrared data link between

the programming unit and the display, program-

ming of the magicball is simplified and allows

customizing. In addition, the life-span of the

product is increased while running costs de-

crease.

IR Data Link: Remote Control => MagicballThe data transmission via in-

frared remote control unit is uni-

directional from the programming

unit to the display. The infrared

signal of the remote control is bi-

phase coded , similar to the stan-

dard RC 5 code, and consists of

a pre-signal and the main signal

[1]. A photodetector - with an in-

tegrated preamplifier, a demodu-

lator and a filter - is used for de-

tecting the infrared signal of the

remote control unit [2]. Since the

receiver module is placed on the

3 RESEARCH38

Fig. 1: Side and top view of the magicball-display-system

T

rotating parts of the magicball,

an optical system (e.g. a special

mirror) is necessary to ensure an

uninterrupted connection be-

tween transmitter and receiver.

This is why a dynamically bal-

anced body made of polymethyl-

methacrylate (PMMA) has been

developed.

Fig. 2 shows the path of an ex-

emplary infrared ray from the re-

mote control unit through the

PMMA-body onto the detector. The loss of in-

tensity caused by the reflection and transmis-

sion involving the PMMA-body is less than 10%.

Tests show that the configuration consisting of

the remote control unit, PMMA-body and pho-

todetector enables a transmission range of up

to 10 m. In addition to the hardware components,

specific software has been developed to pro-

cess the incoming infrared bi-phase coded sig-

nal. After sampling the signal the microcontrol-

ler reassembles the keycode of the remote

control unit and displays the new character on

the surface of the magicball.

IR Data Link: Notebook <=> MagicballFor the data transmission to the magicball

and vice versa , the notebook-PC needs an in-

frared interface (built-in or additional) which op-

erates according to the SIR (Serial Infrared) stan-

dard of the IrDA (Infrared Data Association) [3].

The SIR standard enables a data transmission

range of up to 3 m, at angles of up to 15° and at

a data rate of 115.2 kbit/s. For notebook-PCs

without a built-in infrared interface a dongle fit-

ting to the serial port of a PC has been designed.

The encoder/decoder chip and the transceiver

module with an additional IR-LED for wider trans-

mission ranges support the infrared data trans-

mission based on the SIR standard. The con-

verters allow a power supply of the dongle with

the serial RS232-port. The principle block dia-

gram of the dongle is shown in Fig.3.

In contrast to the remote control unit configu-

ration, the notebook - magicball infrared data

link is bidirectional. Because of the bidirection-

ality and the lower radiation intensity using the

SIR standard, the PMMA-body cannot be used

here. Therefore, a simple transmitter-receiver

system has been chosen, omitting the optical

system used before. This configuration is illus-


15 m

m

15 mm

10 mm

IR-Detektor

epoxy glue

IR-beam

PMMA-body

remotecontrol

Fig. 2: Exemplary infrared signal beam

Fig. 3: Block diagramm of the developed infrared dongle

trated in Fig.4 . Because of the symmetry of the

infrared data link, transmitter and receiver as

shown on the illustration are interchangable. The

bi-phased character of the data link remains in

place.

Due to the rotation of the detector and be-

cause of the angle of detection/emission, we

obtain a periodically recurring optical contact

between transmitter and receiver. Because of a

rotation of about 3600 rpm and a detection an-

gle of 15° the optical contact time (time window)

is nearly 1 ms, while the non-contact time is 15

ms. The software we have developed takes ad-

vantage of the resulting time window, thus en-

abling the data exchange between the notebook

and magicball. The software program first par-

titions the information into blocks of data with

equal size depending on the time window. After

being received, the single data blocks are reas-

sembled. The data security of the transmission

is safeguarded by parity bits on the one hand

and by feedback of magicball on the other hand

( acknowledge / non-acknowledge). For exam-

ple, a bad data block effects a non-acknowledge

response and the block will be transmitted again.

The software for this type of data transmission

and the infrared dongle have been successfully

tested for functionality.

ConclusionIn this report, two

methods of the

wireless program-

ming of a mag-

icball via infrared

data transmission

are discussed. In

the first instance,

programming is

done with an IR remote control using a PMMA

body with integrated photodetector as an opti-

cal medium on top of the rotating arm. In the

second instance, a customized program has

been developed to enable a bidirectional IR data

link between a notebook-PC and the magicball

without an additional body. Many tests have

shown the possibilty of infrared data transmis-

sion on such a rotating receiver system.

In terms of practicability, the remote control

allows a wider transmission range and a lower

level of noise interference. On the other hand

the comfortable editor for the notebook-PC is a

big advantage for the second solution, especially

if large ammounts of text has to be transmitted.

AcknowledgementWe would like to thank the LUMINO Licht Ele-

ktronik GmbH for the possibilty of working on

the display system magicball and the support

during this work.

References[1] Adaptive Micro Systems Infrared Communica-

tion Theory of Operations Abstract, 13.06.88

[2] TEMIC Semiconductors TFMx IR Detector

Photomodules Design Guide, June 1996

[3] St. Williams, I. Millar The IrDA Platform HP

Labaratories, Bristol, 1996

3 RESEARCH40

Sendewinkel

receiver path

receiver

transmitter

detection

angle

emission angle

reception area

Fig. 4: Transmitter - rotating receiver - configuration

3.2.5 Nonlinear hybrid GaAs/AlGaAsmultilayer-heterostructures for high-speed information processing

C. KAMPERMANN, A. KREUDER, AND S. REDLICH

n this report we present theoretical and

experimental results on the nonlinear

optical, electrical, electrooptical and opto-

electronic properties of hybrid GaAs/AlGaAs

multilayer heterostructures. These struc-

tures exhibit fast nonlinear properties and

high sensitivity which can be used for high-

speed information processing in microwave-

photonics.

IntroductionIn recent years optical nonlinearity and bista-

bility in multilayer heterostructures (MLHS) have

received increasing attention because of their

potential use in all-optical high-speed informa-

tion processing systems. Applications are fore-

seen in the areas of photodetectors and modu-

lators with internal amplification as well as fast

optical switching and memory devices. In 1991

He et.al. [3] achieved all-optical bistability in a

30 period GaAs/AlAs structure at an optical in-

tensity of 10kW/cm² [1]. Switching intensities in

the range of kW/cm², however, are orders of

magnitude too high for optical information pro-

cessing. In the presence of an applied electric

field perpendicular to the layers of a MLHS, the

Franz-Keldysh effect together with the accumu-

lation of photocarriers in the GaAs layers and

the voltage dependence of the current through

the device are used to decrease the switching

intensities by five orders of magnitude [2]. These

kind of hybrid MLHS exhibit the lowest switch-

ing intensities in comparison to other device

concepts. Appropriate designed MLHS also ex-

hibit s-shaped negativ differential conductivity

(SNDC), based on bistability between tunneling

and thermionic emission across the heterobar-

riers. Calculations and experiments have shown

that selfsustained voltage oscillations up to 100

GHz occur, if the MLHS is driven in an external

resonator. In this report we present theoretical

and experimental results concerning these novel

kind of devices.

Device StructureFig. 1 shows the cross section of the device

containing a periodical GaAs/ Al0.45Ga0.55As

MLHS. The MLHS consists of 20 bilayers with

nominal thicknesses of 58 nm (GaAs) and 69

nm (AlGaAs). The layers are grown by usual


Fig. 1: Sketch of the device

I

MBE on s.i. GaAs substrates. A 1000nm n+-

GaAs contact layer is introduced for sufficient

high values of the bulk conductance. A number

of different process techniques such as wet etch-

ing and evaporation are used to form the de-

vice. Conventional photolithography is applied

to pattern structures on the wafer. The trans-

mission line is made of a TiPtAu multilayer met-

alization, which is applied to the wafer by evap-

oration. To prevent a short circuit between the

center conductor of the transmission line and

the contact layer at the bottom of the MLHS the

edges of the mesas are coated with polyimide.

A voltage can be applied to the coplanar

transmission line to get a high electric field con-

centrated in the MLHS. With coplanar transmis-

sion lines used as electrical contacts, operation

up to millimeterwave frequencies is possible.

The optical input and output ports are defined

by a via hole in the center conductor of the trans-

mission line.

TheoryTo simulate the device it is necessary to anal-

yse the optical wave propagation as well as the

transport and the accumulation of charge carri-

ers in the MLHS. Additionally, the electrooptical

and the optoelectronic interactions between the

optical and the electrical subsystems, namely

the Franz-Keldysh effect and the generation of

photocarriers in the GaAs layers have to be con-

sidered.

Optical propertiesFor an optical wave propagating in a period-

ically layered medium like a MLHS it has been

shown that, there exists an optical resonance

effect when the optical wavelength is close to

the optical stopband. This means that the inten-

sity distribution in the MLHS depends on the

optical wavelength and strongly increases when

approaching the resonance. This effect is es-

sential for the behaviour of the device and we

had to take this into consideration for our simu-

lations. In the linear case one can use the trans-

fer-matrix method (TMM) to calculate the inten-

sity distribution in the layered medium. In the

nonlinear case of the MLHS the standard TMM

cannot be applied because of the intensity de-

pendent refractive index of the GaAs layers. To

overcome this problem we used a generalised

3 RESEARCH42

Fig. 2: (a) Linear reflectivity spectrum of a InGaAlAs/InAlAs MLHS and average optical intensity in the

InGaAlAs layers of the structure.( b) Reflectivity over incident optical intensity of the same MLHS for

two different wavelength.

form of the TMM [3] where the GaAs layers are

divided into a number of sublayers. Assuming

that the optical intensity in each sublayer is con-

stant one can determine the intensity dependent

refractive indices by using the boundary condi-

tions of the wave amplitudes in two adjacent lay-

ers. Then, specifying the field at the end of the

structure one can apply the standard TMM to

calculate the nonlinear characteristics of the

MLHS. Figure 2(a) shows the linear reflectivity

spectrum of a MLHS consisting of 40 pairs of

InGaAlAs/InAlAs bilayers, lattice matched to InP.

As can be seen at a glance, by using the In-

GaAlAs/InP system one can shift the operation

wavelength of the device towards 1.5µm, an

important wavelength for applications in optical

communication systems. Like demonstrated for

the AlGaAs/GaAs system by He et.al. all-opti-

cal bistability based on the nonlinear refractive

coefficient n2 without an applied electric field can

also be observed. The nonlinear reflectivity-ver-

sus-intensity characteristics of the structure,

calculated with the above method, are shown in

Fig.2(b). The curves are calculated for two dif-

ferent wavelengths at the long-wavelength side

of the stopband. Both are Z-shaped and exhibit

a bistable hysteresis loop. As mentioned above

we have experimentally shown that by applying

an electric field perpendicular to the layers of

the MLHS switching intensities below 10mW/cm²

can be achieved. Therefore, besides the optical

properties, the electronic, optoelectronic and

electrooptical properties of the MLHS are from

special interest.

Electrical propertiesOur model of the transport and the accumu-

lation of charge carriers in the MLHS is based

on an analytical model of a heterostructure hot

electron diode (HHED) by Wacker et.al.[4]. The

HHED shows S-shaped negative differential re-

sistance (NDR) or differential gain and consists

of two undoped adjacent heterolayers (GaAs/

AlGaAs) with ohmic contacts. In this structure

two conduction mechanisms exist: At low fields

the current is limited by tunneling through the

AlGaAs layer (low conductance state on

theFig.3(a)). At higher fields the charge carriers

are heated up to sufficiently high energies so

that thermionic emission over the barrier be-

comes dominant (high conductance state on the

Fig. 3(b)). The extremely fast transition between

these conduction modes leads to NDR or differ-

ential gain. We extended the model of Wacker

et.al. to calculate the electronic properties of

MLHS. The physical processes of the charge

transport in the MLHS are sketched in Fig. 4(a).

As an additional effect, the cooling of the charge

carriers, which means the capture by the GaAs

wells is included. The numerically obtained cur-

rent-density-voltage characteristics (see Fig.

4(b)) are in good agreement with the results of

Monte-Carlo simulations published by Reklaitis

[5]. Fig. 4(b) further elucidates a pronounced S-

shaped NDR for (GaAs 100nm / AlGaAs 70nm)

and for thicknesses used in our device struc-

ture merely a preliminary stage of NDR.


GaAs AlGaAs

W

GaAs AlGaAs

W

b)a)

Fig. 3: Schematic conduction band structure

of a GaAs/AlGaAs heterostructure with a

perpendicular electric field. The two possible

conduction states are shown.

InteractionFig.5 shows schematically the system of a

hybrid MLHS [5]. The optical and the electrical

subsystems are coupled by two interaction

mechanisms, the generation of photocarriers

(optoelectronic) and the Franz-Keldysh effect

(electrooptical). The photocarriers are generat-

ed by optical absorption in the GaAs layers. Due

to the nonlinear electrical properties of the MLHS

a small photocurrent leads to a strong variation

of the internal voltage distribution and, by the

Franz-Keldysh effect (FKE), to a large change

of the refractive indi-

ces in the GaAs lay-

ers. It has been shown

that the same change

of the refractive index

can be achieved at

much lower optical in-

tensities as for the in-

trinsic optical nonlin-

earity. Including these

mechanisms the feed-

back, which deter-

mines the optical bi-

stability of the hybrid MLHS can be described

as follows: An incident optical intensity leads to

an optical intensity in the MLHS, where a part of

the light will be absorped. The generated carri-

ers give rise to a photocurrent. This in turn leads

to a change of the voltage drop across the GaAs

layers and a change of the refractive indices of

this material. This variation finally changes the

reflectivity of the hybrid MLHS and in turn the

absorped optical intensity. Thus, a feedback loop

exists. The interaction mechanisms and the

feedback loop are also implemented in our mod-

el of the hybrid MLHS. Thus, the experimentelly

observed device characteristics including the op-

tical, electrooptical, optoelectronic and optically

induced electrical bistability of a hybrid MLHS

could be verified (Fig. 6).

Experimental resultsComprehensive measurements of the opto-

electronic properties of hybrid MLHS have

shown a high optical sensitivity of these devic-

es. At a reverse bias of V=30V we have mea-

sured photocurrents of around I=1mA and dark

currents of merely 20nA (see Fig. 7).

A bias voltage also changes the reflectivity,

as shown on theFig. 8(a) , where the electroop-

3 RESEARCH44

CoolingEmission

Heating

WC

TOTALCURRENT

2,0 3,0 4,0 5,0 6,0

0

400

800

1200

1600

curr

ent

dens

ity in

kA

/c

voltage in Va) b)

GaAs 100nm / Al0.45Ga0.55As 70nmGaAs 58nm / Al0.45Ga0.55As 69nm

Fig. 4: (a) Schematic view of the conduction band structure of a MLHS with

a perpendicular electric field and the physical processes of the charge

transport. (b) Current density-vs.-voltage characteristic of a heterostructure

for two different layer thicknesses.

ZV

i

0

0

Fig. 5: Cross section of the MLHS with I the

current flow and P the optical wave propagat-

ing through the device.


Fig. 6: (a) Measured current-voltage characteristic of a hybrid MLHS under

illumination. (b) Calculated I-V curve of the same device and operation pa-

rameters.

voltage in V

-30 -20 -10 0 10 20 30

10-8

10-710-610-510

-410

-310

-2dark

curr

ent i

n A Popt = 1mW

Fig. 7: Measured current-voltage character-

istics in the dark and illuminated case.

cut-off frequencies of up to 420 MHz, measured

from MLHS devices based on another contact

geometry. The frequency response of these de-

vices is RC limited, therefore higher cut-off fre-

quencies should be reached by using the trans-

mission line design. The coupling of the

electrooptical modulation and the optoelectron-

ic properties of the device leads to optical bista-

bility, which has been measured at optical in-

tensities below 10mW/cm² (Fig. 9).

ConclusionIn this report new theoretical and experimen-

tal results concerning hybrid MLHS are present-

-8

-4

0

4

105 106 107 108

frequency in Hz

A = (600µm)²

rel.

mod

ulat

ion

in d

B

wavelength in nm860 880 900

-2

0

2

4

6 -10 V -20 V

mod

ulat

ion

cont

rast

in

Fig. 8: (a) Modulation contrast over wavelength at different reverse biases. (b) Relative modulation

characteristic as a function of frequency.

tical modulation

near the band-

gap wavelength

is due to the

Franz-Keldysh

effect. A modula-

tion contrast of

about 6dB could

be reached at a

voltage change

of 20V. Time and

frequency do-

main measure-

ments were carried out to investigate the dynamic

properties of the MLHS. We have determined

3 RESEARCH46

above 1 MHz are to be expected. In compari-

son, the results differ only slightly from those

obtained with commercially available superlu-

minescence diodes.

64 Channel Silicon Driver CircuitThe TTL-compatible silicon chip was de-

signed to power the above mentioned array of

8 x 8 LEDs, with each LED driven independent-

ly by an output current adjustable in the range

of 0 - 10 mA. Based on the requirement of a

maximum input current for the IC of 1 mA, a

current amplification with a factor of 10 must be

achieved. The realized circuit consists of the

following main components: (i) two 3 to 8 de-

multiplexers for binary coded x- and y-selection

of each LED driver, (ii) a current mirror to tune

the gain of 10 and to shorten the input Iin if Izero

is set to zero, and (iii) 64 independently select-

able LED drivers containing xy-selection units

and a capacitor providing a constant output cur-

rent Iout during regeneration cycles. In Fig. 3 the

layout of the chip, consisting of 64 identical

amplifier cells, bondpads, and demultiplexing

circuits, is sketched. Each cell requires an area

of 200 * 200 µm², leading to a total square sur-

face of 1.6 * 1.6 mm² and consequently to a pixel

density of 127 DPI. Together with control cir-

cuitry and pads for external wire bonding, a to-

tal chip dimension of only 2.3 * 1.6 mm² is

achieved.

The electrical characterization of the silicon

circuit by measuring the pulse response with a

sampling oscilloscope was leading to rise and

fall times less than 250 ns. This results in a cut-

off frequency of fc ³ 1.45 MHz. Due to the fact

that the I-V characteristic referred to the input of

the circuit shows strong non-linear behaviour,

an interface circuit (voltage driven current

source) was applied, leading to a

decrease of the cut-off frequen-

cy. Together with a D/A convert-

er computer board the system

shown in Fig. 4 was built, provid-

ing a good linearity between in-

put voltage and output current.

Hybrid integration In Fig. 5 the final device con-

sisting of the silicon driver IC

bonded to the LED array is de-

picted. A composition of almost

Fig. 2: Photograph of the array with one LED

illuminated .

Fig. 3: Layout of silicon integrated circuit with detail of LED

drivers.

3 RESEARCH48

eutectic solder (60 wt % Sn, 40 wt % Pb) is evap-

orated onto the metal contacts of the LED ar-

ray. After the reflow process, at

200° C for 10 seconds, both the

array and silicon driver IC are ad-

justed and bonded together.

Since the PbSn layer thickness is

much smaller than electroplated

PbSn due to the evaporation pro-

cess, this bonding technique is a

mixture of thermocompression

and soldering. Following the flip-

chip process the ground contact

for the LED array, together with

connections for packaging of the

Fig. 4: Computer controlled silicon driver circuit.

Fig. 5: Cross-sectional view of the silicon

driver circuit bonded to the LED array.

photonic IC, are established using

wire bonding technique.

ApplicationsWith this system presented

here several applications can be

realized. One example is a spe-

cial kind of vision aid for blind

persons with a blurred cornea. In

Fig. 6 one possible realization of

this vision aid is sketched. Under

various circumstances (accidents

where the cornea is damaged, e.g

in explosions or by erosion due to acid) a num-

ber of people loose their sight although their ret-

ina is fully intact. A photodetector array converts

images into digital information wirelessly trans-

mitted to a miniature display like our proposed

model, implanted into the lens. This display

projects a very simple image onto the retina, of-

fering a primitive vision.

ConclusionExperimental investigations of both the silicon

circuit and the LED array show cut-off frequen-

Fig. 6: Vision aid for people with blurred cornea.


cies beyond 1 MHz, leading to the conclusion that

this hybrid integrated circuit is a promising sub-

system not only for parallel optical information

processing systems but also for a novel applica-

tion of photonic integrated circuits in the field of

neurotechnology.

AcknowledgementsThis work was financially supported by the

Federal Ministry for Education, Science, Re-

search, and Technology (BMBF) in the frame of

the EPI-RET: Retina Implant project under

contract number 01 IN 501 G. The authors would

like to thank G. Sixt (TEMIC Telefunken, Heil-

bronn) for providing the GaAsP/GaP wafer and

R. Klinke (Fraunhofer Institut-IMS, Duisburg) for

helping designing the silicon chip. Thanks goes

also to K. Heimann (Uni-Augenklinik, Köln) for

giving an insight into several ophtalmological

problems.

References[1] H.F. Bare et al., IEEE Photon. Technol. Lett., 5,

2, pp.172, 93

[2] G.W. Turner et al., IEEE Photon. Technol. Lett.,

3, 8, pp.761, 91

[3] A.J. Moseley et al., Electron. Lett., 27, 17,

pp.1566, 91

[4] K. Werner, IEEE Spectrum, pp.30-39, Jul. 94

[5] W.R. Imler, et al., IEEE Trans. Compon., Hybr.

Manufact. Technol., 15, 6, pp.977, 92

[6] Y. Nitta et al., IEEE Photon. Technol. Lett., 4, 3,

pp.247, 92

[7] H. Yonezu et al., Electron. Lett., 25, 10, pp.670,

89

[8] M.A. Brooke et al., Optics & Photonics News,

pp.26, Jun. 93

[9] M. Wale et al., IEEE Circuits & Devices, pp.25,

Nov. 92

3 RESEARCH50

DD

DD

Fig. 1: Monolithic NLTL with periodic array of

Schottky diodes

C(V)

L/2 R/2

V

L/2R/2Ik

Gk

Fig. 2: Equivalent circuit for one element of

the NLTL

3.3 Millimeterwave Electronics

3.3.1 Picosecond pulse generationon monolithic nonlinear transmis-sion lines using high-speed InP-HFET diodes

R. HÜLSEWEDE

lectrical pulses with transients less

than 5 ps are generated and com-

pressed on monolithic InP-HFET diode non-

linear transmission lines. The transients are

measured by time domain electro-optic sam-

pling technique and the waveforms show

good agreement with numerical results. Ad-

ditionally, frequency domain measurements

and numerical simulations reveal that the

nonlinearities work with frequencies higher

than 400GHz for 20µm x 20µm InP-HFET di-

odes. Instead of costly ion implantation tech-

nology a chemical recess is used to isolate

the active structures.

IntroductionMonolithic nonlinear transmission lines

(NLTL) are circuits with an alternating arrange-

ment of coplanar waveguides and Schottky di-

odes as shown in Fig. 1.

The capacitance-voltage characteristic of the

Schottky diodes in combination with the low pass

filter characteristic of the periodic structure leads

to the generation of shock waves and the for-

mation of pulses with (sub-) picosecond tran-

sients [1,2]. Therefore, these circuits are impor-

tant for novel measurement and characterization

methods for new high-speed devices.

For numerical simulations the equivalent cir-

cuit as shown in Fig.2 is used leading to a dif-

ference equation for current and voltage at each

element of the NLTL.

Applying a transition to a differential equa-

tion one obtains the following wave equation

which considers separately the influences of the

nonlinearity of the diodes, the periodic structure

and the losses of the transmission line (for de-

tails see [3,4]):

∂∂

∂∂

Vx

C VC

Vt= −

+1

0

( )

∂∂

∂∂

L C Vt

RL V

CG

Vt

+⋅

− +120

3

30

2

2 (1)

Here the nonlinearity of the diodes is consid-

ered by the normalized capacitance-voltage

dependence C(V)/C0 , where C0 is the capaci-

tance at the operating point of InP-HFET diode

In order to improve the nonlinearity of the

Schottky diodes in NLTLs d-doped diodes based

3.3 Millimeterwave Electronics 51

E

on InP-HFET layer structures are used ( see

Fig.3 and [5]). High electron concentration in the

d-doped layer (4.9 1012cm-2), maximum mobili-

ty (10900 Vs/cm-2), and the 2 dimensional elec-

tron gas (2-DEG) in the InGaAs channel are

special features of this layer structure at T=300K.

The strong nonlinearity of the HFET-layer struc-

ture is shown in Fig. 4 where the normalized ca-

pacitance-voltage characteristic of an InP-HFET

diode is sketched. Over the 0.5V bias range

around the working-point a 2200% change of

the capacitance is achieved. This is a 20x great-

er nonlinearity than d-doped GaAs Schottky di-

odes used in NLTLs described in [6]. One rea-

son for this strong nonlinear behaviour is the

depletion of the 2-DEG underneath the nega-

tive biased Schottky contact. Using InP-HFET

diodes in NLTLs the nonlinear interaction of the

propagating waves is increased and thus the

line-losses are decreased due to shortening of

the line length. Another advantage is the appli-

cation of a C4H6O4,H2O2,NH3 recess [7] for elec-

trical isolation of the InP-HFET diodes in NLTLs.

Thus, no costly ion implantation process is need-

ed and no preparation of an accelerator is re-

quired.

InP-HFET NLTLIn a first step a 10 diode periodic InP-HFET

NLTL was fabricated in order to verify experi-

mental and numerical results. For that purpose

frequency domain measurements along the cen-

ter conductor are shown in Fig. 5 (see also [8]

and P. Bussek et al., Time- and frequency do-

main electro-optic field mapping of nonlinear

transmission lines, in this annual report).The

electro-optic signal of the excited input wave

(15GHz, 25dBm) decreases from input to out-

put of the NLTL, whereas the generated har-

monic signals at 30GHz, 45GHz and 60GHz in-

crease. Using the nonlinearity of InP-HFET

diodes (Fig.4) and a FFT the nonlinear wave

propagation on this NLTL is simulated. The re-

sult is shown by the grey lines in Fig. 5. With

respect to the -128dB noise level and the +/-5dB

accuracy of the sampling signal both results are

in good agreement. This agreement and the

numerical value for C0/G = 3,5 10-13s indicates

that the nonlinearity works with frequencies high-

er than 400GHz for the 20µm ´ 20µm InP-HFET

diodes.

Thereupon different NLTLs are simulated based

on these perceptions in order to generate one

Schottky contactOhm contactInGaAs-channelInAlAsInGaAlAsInPδ-Si

Fig. 3: Schematic profile of an InP-HFET

diode (for details see [5])

0-1-2-3

10

1

0.1

0.01

Bias voltage(V)

C(V

)/C

o

Fig. 4: Normalized capacitance-voltage

characteristic of InP-HFET diodes

3 RESEARCH52

single pulse per period of the sinusoidal input sig-

nal. The simulation in Fig.6a demonstrates the

generation and compression of single pulses out

of the exiting input signal (6.5GHz, 2.5V) on a

graded NLTL with increasing values of L and C0

in direction of the propagating microwave. The

transient with minimum 10-90% fall time of 4ps

and an amplitude of 1.8V is shown in Fig. 6b.

After fabrication of this graded InP-

HFET NLTL using self aligned op-

tical contact lithography process-

es [7] the electro-optic sampling

set-up was modified making time

domain measurements (see P.

Bussek et al., Time- and frequen-

cy domain electro-optic field map-

ping of nonlinear transmission

lines, in this annual report). In

Fig. 7 a top view of the graded

InP-HFET NLTL is figured includ-

ing the four points of measurement

(a)-(d). The frequency of the input

signal is 6.5GHz with an amplitude

of 3.5V (measured at 50W). Clear-

ly the steeping of a shock wave

(k=11) and the generation of a sin-

gle pulse (k=31) with FWHM of

10ps and a fall time of about 5ps

is observed. Thus, picosecond

pulse generation on NLTL using

high speed InP-HFET diodes is

shown for the first time. Addition-

ally, the numerical results (Fig.6)

at the corresponding points of

measurements are plotted in Fig. 7

(grey lines). The agreement of

waveforms is satisfying demon-

strating that the fundamental

mechanisms of nonlinear wave

propagation on NLTLs have been considered in

equation (1).

ConclusionIn this work the generation and compression

of picosecond pulses on InP-HFET NLTL is dem-

onstrated. Advances have been achieved by

applying high speed InP-HFET diodes exhibit-

0 500 1000 1500 2000 2500

x (µm)

Sig

nal (

dBm

)

-90

-100

-110

-120

-130

-140

-80

Input signal: 15GHz,27dBmexperimentsimulation

15GHz

30GHz

45GHz

60GHz

Fig. 5: Generation of harmonic signals on periodic InP-HFET

NLTL. The structure of NLTL is sketched at top of this figure.

(Data of simulation: L pH=120 , C0 = 1.6 pF,

R L s/ = ⋅ −2 1010 1, C G s0133 5 10/ .= ⋅ − )

Signal voltage

x

t

1V

10ps(a)

Sig

nal v

olta

ge

(V)

Time (ps)

1

-2

0

-1

0 40 12080

(b)

(b)

Fig. 6: Simulation of pulse generation on a graded InP-HFET

NLTL; (a) development of a sinusoidal signal along the NLTL,

(b) transient with minimum fall time (Data of simulation:

L pH= 960 , C0 = 1.76 pF, R L s/ = ⋅ −2 1010 1,

C G s0133 5 10/ .= ⋅ − , grading: 0 89. αx , α = ⋅ −8 4 1010 1. s )


ing strong nonlinearities. The numerical simula-

tion has been improved by considering sepa-

rately the influence of nonlinearity, periodic struc-

ture and losses to nonlinear wave propagation

on NLTLs.

AcknowledgementThe author would like to thank Dipl.-Phys.

U. Auer (Fachgebiet Halbleitertechnik / -technol-

ogie) for processing of the transmission lines and

Dr. D. v.d.Weide (at that time: Max-Planck-In-

stitut für Festkörperforschung, Stuttgart) for lend-

ing a suitable mask for optical contact lithogra-

phy processes.

References[1] M.J.W. Rodwell et al, Active and nonlinear wave

propagation devices in ultrafast electronics and

optoelectronics, Proc. IEEE, Vol. 82, No. 7, 1994,

pp. 1037-1059

[2] D. Jäger, Characteristics of travelling waves

along nonlinear transmission lines for monolithic

integrated circuits: A review, Int. J. Electron.,

Vol. 58, 1985, pp. 649-669

[3] D. Jäger, Pulse generation and compression on

nonlinear transmission lines, workshop on Pi-

cosecond and Femtosecond Electromagnetic

Pulses: Analysis and Applications, MTT-S Symp.

Dig., 1993, pp. 37-57

[4] R. Hülsewede et al, CAD of pulse compression

on nonlinear transmission lines, Proc. MIOP 95,

Sindelfingen, 1995, pp. 511-515

[5] U. Auer et al, InP based HFETs with high qual-

ity short period InAlAs/InGaAs Superlattice

Channel Layers, J. o. Crystal growth, vol. 146,

1995

simulationexperiment

(a)

(d)(c)(b)

(a)(b)

(c)

(d)

k = 31

k = 7

k = 15k = 11

input

output time (ps)

sign

al v

olta

ge (

a.u.

)

-80 -40 0 40

-80 -40 0 40si

gnal

vol

tage

(a.u

.)

sign

al v

olta

ge (

a.u.

)

sign

al v

olta

ge (a

.u.)

-80 -40 0 40-80 -40 0 40time (ps)time (ps)time (ps)

Fig. 7: Pulse compression on graded InP-HFET NLTL (top view of the processed NLTL in the upper

left side of this figure, (a)-(d): points of electro-optic measurements)

3 RESEARCH54

[6] D.W. van der Weide, Delta-doped Schottky di-

ode nonlinear trans-mission lines for 480-fs, 3.5-

V transients, Appl. Phys. Lett. Vol. 65, No. 7,

1994, pp.881-883

[7] C. Heedt et al, On the Optimisation and Reli-

ability of Ohmic- and Schottky Contacts to InAlAs/

InGaAs HFET, Proc. 4th InP & Related Materi-

als Conference, Newport, USA, 1992

[8] Report on the Special Collaborative Programm

SFB 254, 1993-1995, Gerhard-Mercator-

Universität - GH - Duisburg, 1995

3.3.2 Millimeter wave power genera-tion on nonlinear transmission lines

R. HÜLSEWEDE, V. K. MEZENTSEV, AND

I. V. RYJENKOVA

n this paper nonlinear transmission

lines are described which are used to

generate millimeterwave signals with high

efficiencies. In particular, arrays of monolith-

ic varactor diodes loading a coplanar

waveguide are studied which can be applied

for travelling wave harmonic generation

where special phase matching and filter

structures give rise to high conversion effi-

ciencies. A second transmission line consist-

ing of any array of resonant tunneling diodes

is used as a distributed active device which

can generate millimeterwave power at fre-

quencies as determined by a resonance con-

dition of the resonator structure under study.

In this paper theoretical and numerical re-

sults are presented based upon experimen-

tal data.

Introduction:The generation of millimeter waves by har-

monic frequency generation and active wave

propagation along nonlinear transmission lines

(NLTLs) has recently become a subject of ma-

jor research activities [1-4]. However, the pow-

er efficiencies achieved so far are small because

millimeterwave power is converted into undes-

ired frequency components when the dispersion

and filter characteristics of the NLTL are not

designed in a suitable way. In this paper, firstly

we describe the bi-modal NLTL which uses con-

cepts of nonlinear optics aiming towards achiev-

ing phase matching condition between the fre-

quency components under study [5,6]. In

particular, we study a bi-modal NLTL where, as

an example, the phase velocity of the second or

third harmonic equals that of the fundamental

wave and where other components are sup-

pressed by a suitable filter structure leading to

a distinct cut-off frequency [7-9]. Secondly, we

discuss the characteristics of a travelling-wave

tunneling-diode transmission line resonator ca-

pable of generating high power millimeter wave

signals [7-9].

In Fig. 1, the basic structure of an NLTL is

sketched consisting of a suitable array of non-

linear devices D in a passive coplanar wave-

guide [2,3]. As nonlinear elements, we have

studied Schottky diodes, quantum barrier var-

DD

DD

Fig. 1: Sketch of a nonlinear transmission

line


I

actor structures (QBV), as well as resonant tun-

neling diodes (RTDs).

The second characteristic feature of the cir-

cuit in Fig. 1 is the dispersion which is mainly

determined by the arrangement of the diodes,

which can be periodic, bi-periodic, graded etc.

The dispersion itself controls the phase veloci-

ties of different spectral components and hence

the strength of interaction and the superposi-

tion in the time domain. Reflections at input and

output ports further determine the resonance

behavior of the whole structure.

Harmonic frequency genera-tion:

We have studied nonlinear

wave propagation along NLTLs of

Fig. 2. In a first example the pa-

rameters used are those of an

experimental device on InP-HFET

substrate as discussed in [10].

Input frequency and power are

100GHz and 11dBm as delivered

into a small-signal characteristic

impedance of 50W. As a numeri-

cal tool we have used a continu-

um approximation on the basis of

a corresponding nonlinear evolu-

tion equation as described in

[10,11] and compared the results

with a CAD model based on a dis-

crete representation of the NLTL

, cf.[10].

The results of our numerical

calculations are plotted in Fig. 3

showing the spatial distributions

of the amplitudes of the funda-

mental and second harmonic

wave. As can be seen, the ampli-

tudes of the two waves at the in-

put are comparable, which leads to power effi-

ciencies > 70% for second harmonic genera-

tion (SHG), here at 200GHz.

In a second numerical experiment we have

studied third harmonic generation (THG) on a

bi-modal NLTL of Fig. 2(b) assuming special

quantum barrier varactor diodes [12] with a sym-

metric capacitance voltage relationship. Fig. 4

presents the numerical results. Note that in this

case of THG phase matching is achieved be-

tween frequencies f1 = 72GHz and f3 = 216GHz.

As can be seen from Fig. 4, the 72GHz input

signal is converted into millimeter wave power

RTDRTD

RTDRTD

InP 350 µm

InGaAs 40 nm

InGaAs 500 nm n+

InGaAs 4.3 nm

InGaAs 40 nm undoped

InGaAs 500 nm n+

undoped

undoped

undoped

undoped7.2 nmInAlAs

InAlAs 7.2 nm

o

o

C

CC

CInP 350 µm

InGaAs 250 nm

InGaAs 500 nm n+

InGaAs 25 nm

InGaAs 25 nm

InGaAs 250 nm n

InGaAs 250 nm n+

undoped

undoped

n

InAlAs 25 nm undoped

o

o

C

CC

C

InP 350 µm

InGaAs

300 nm

InGaAs

50 nm n

n

n

n50 nm

InAlAs

800 nm

-

-

+

+

InGaAs

(a)

(b)

(c)

Fig. 2: Monolithic NLTL on InP substrate for millimeter wave

generation. (a) Bi-modal NLTL with Schottky varactor diodes for

SHG, (b) bi-modal NLTL with QBV for THG, and (c) RTD-NLTL

for a distributed oscillator.

3 RESEARCH56

at 216GHz with an efficiency of about 25%. Again,

the third harmonic is available at the input of the

NLTL because the propagation characteristic is

that of a backward wave [5,6].

Tunneling diode NLTL:Very recently, another type of NLTL has be-

come known where resonant tunneling diodes

are used as nonlinear elements [3]. However,

such RTD-NLTLs can also be used for nonlin-

ear active wave propagation ef-

fects leading to a travelling wave

oscillator, when a transmission

line with limited length, provided

by short circuits at input and out-

put ports, for example, is used

[7,8]. In a numerical experiment

we have studied the generation

of millimeter waves in a RTD-

NLTL resonator. The results are

plotted in Fig.5 revealing self-gen-

erated oscillation at 170 GHz af-

ter about 700 ps which is the

switch-on time.

In a further numerical example

we have studied the relationship

between the length N of the os-

cillator and the frequency f(n, N) of the self-gen-

erated oscillations where n = 1, 2, , N defines

the mode. Fig. 6 shows the results where the

dots represent the results of the simulations.

Clearly, a decreasing N leads to an increasing

f(n,N) because the wavelength, as given by

2 x N, decreases. In Fig. 6 a comparison with

analytical results is additionally carried out where

f(n, N) is given by

0

0.2

0.4

0.6

0.8

1

1.2

0 20 40 60 80 100 120 140 160number of elements, k

f1

f = 2f2 1

Fig. 3: Spatial distribution of amplitudes at

frequencies f1 and f2 for the NLTL of Fig.2(a).

0 20 40 60 800

0.1

0.2

0.3

0.4

0.5

0.6 f1

f3

100 120 140 160number of elements, k

Fig. 4: Spatial distribution of amplitudes at

frequencies f1 and f3=3f1 for the NLTL of

Fig.2(b)

0 100 200 300 400 500 600 700 800

-0.4

0

0.4

0.8

1.2

1.6

time, ps

volta

ge, V

0 100 200 300

ampl

itude

400frequency, GHz

Fig. 5: Generation of a 170 GHz signal on a tunneling diode

NLTL. The spectrum is shown in the inset.


f n NLC

nN( , ) sin( )= 1 1

2ππ , n = 1,2...,N

(1)

as calculated from the dispersion relation for a

cascaded LC - chain neglecting losses.

ConclusionIn this paper, specific NLTLs are presented

which are capable to generate millimeter waves

with high conversion efficiencies. The NLTLs are

compact, easily fabricated using standard InP

technology, suitable for monolithic integration,

and can provide high output powers. We there-

fore conclude that the travelling wave concept

under study can provide a solution to the prob-

lem of realizing efficient millimeter wave signal

sources.

References[1] A. Scott, Active and nonlinear wave

propagation in electronics, John Wiley &

Sons, New York, 1970

[2] D. Jäger, Characteristics of travelling

waves along nonlinear transmission lines for

monolithic integrated circuits: A review, Int.

J. Electron. 58, 649-669 (1985) (invited pa-

per)

[3] M.J.W. Rodwell, S.T. Allen, R.Y.Y. Yu,

M.G. Case, U. Bhattacharya, M.Reddy, E.

Carman, M. Kamegawa, Y. Konishi, J. Pusl,

R. Pullela, Active and nonlinear wave propa-

gation devices in ultrafast electronics and op-

toelectronics, IEEE Proc., vol. 82, no. 7, pp.

1037-1059, 1994 (invited paper)

[4] E. Carman, M. Case, M. Kamegawa, R. Yu, K.

Giboney, and M.J.W. Rodwell, V-band and W-

band broadband, monolithic distributed frequency

multipliers, in: 1992 IEEE MTT-S Digest, pp.

819-822, 1992

[5] B. Wedding and D. Jäger, Phase-matched sec-

ond harmonic generation and parametric mixing

on nonlinear transmission lines, Electron. Lett.

17, 76-77 (1981)

[6] D. Jäger, Nonlinear slow-wave propagation on

periodic Schottky coplanar lines, IEEE Micro-

wave and Millimeter-Wave Monolithic Circuits

Symposium, St. Louis 1985, Symp. Dig., 15-17

(1985)

[7] I. V. Ryjenkova, V. K. Mezentsev, S.L.Musher,

S. K. Turitsyn, R. Hülsewede, and D. Jäger, Mil-

limeter Wave Generation on Nonlineat Transmis-

sion Lines, Proc.1996 International Workshop

on Millimeter Waves, April 11-12, Orvieto, Italy,

1996

[8] V. K. Mezentsev, S.L.Musher, I. V. Ryjenkova, S.

K. Turitsyn, R. Hülsewede, D. Jäger, Travelling

wave generation of millimeter waves in bi-modal

N0 5 10 15 20 25 30

0

100

200

300

400

500

600

numerictheory

n = 1

Fig. 7: Oscillation frequency vs. number N of elements

of the RTD-NLTL, theory according to eq. (1)

3 RESEARCH58

NLTLs, Proc. 26th European Microwave Con-

ference 1996, Prague

[9] I. V. Ryjenkova, V. K. Mezentsev, S.L.Musher, S.

K. Turitsyn, R. Hülsewede, and D. Jäger, Milli-

meter Wave Generation on Nonlineat Transmis-

sion Lines, Ann. des Telecomm., Special Issue

(submitted).

[10] R. Hülsewede, U. Effing, I. Wolff, and D. Jäger,

CAD of pulse compression on nonlinear trans-

mission lines , Proc. MIOP 95, Sindelfingen, pp.

511-515

[11] M. Dragoman, R. Kremer, and D. Jäger, Pulse

generation and compression on a travelling-wave

MMIC Schottky diode array, in: Ultra-Wideband,

Short-Pulse Electromagnetics, H.L. Bertoni, L.

Carin, and L.B. Felsen, eds., Plenum Press, New

York, pp. 67-74, 1993

[12] M.A. Frerking, J.R. East, Novel heterojunction

varactors, IEEE Proc., vol. 80, no. 11, pp. 1853-

1860, 1992

3.3.3 Nonlinear RTD circuits forhigh-speed A/D conversion

I. JÄGER

n this report a novel nonlinear MMIC

structure based upon monostable reso-

nant tunneling diodes (RTDs) is studied. For

the first time, it is shown that an input signal

can be converted into a set of output spikes

to be used for GHz A/D conversion.

IntroductionA huge amount of work has recently been

dedicated to the study of resonant tunneling di-

odes (RTDs) which can provide gain and can

directly be used as the key components for os-

cillator circuits approaching the THz frequency

range [1]. The underlying characteristic is a non-

linear N-shaped current voltage relationship even

at millimeterwaves. The lack of these very inter-

esting devices, however, is the low power con-

version efficiency and the small output power

levels [2]. Up to now the only solution to the latter

problem which has become known is the use of

a series i.e. distributed connection of several

RTDs using MMIC technology [3,4]. Such a RTD

nonlinear transmission line (NLTL) can further

provide the basis of very interesting microwave

signal processing devices as has been predicted

by Crane already in 1962 [5].

In this paper, we discuss first the fundamental

concept of nonlinear active wave propagation

effects along monostable RTD-NLTLs utilized to

generate a set of spikes from anelectrical input.

The idea of such a transmission line, where loss-

es are exactly compensated by distributed am-

plification, dates back to the so called neuristor´

[5] as a line-analog of axons in the nervous sys-

tem, where the information of an input signal is

converted into a number of output spikes, travel-

ling in a stationary way for arbitrarily long dis-

tances. In a second step, we describe an electri-

cal circuit, where the monostable RTD-NLTL is

the main building block to realize a n-bit A/D con-

RTD

L GRTD

L G

Fig. 1: Sketch of a nonlinear array of

monostable resonant tunneling diodes in a

coplanar transmission line


I

verter at GHz rates, similar to the lumped RTD

A/D converter described in [7-9].

RTD-circuitThe array of monostable RTDs is sketched

in Fig.1. One can see a coplanar transmission

line which is periodically loaded, cf.[3,4], with

RTDs shunted by LG circuits -here air bridges- in

order to provide monostable behavior, see [6].

The cross section of the MMIC structure in Fig. 1

has been described in [3,4].

The simulation carried out in this paper is

based upon a suitable equivalent circuit, as

shown in Fig.2. Each section consists of an T-

equivalent representation of the transmission

line. The nonlinear element in Fig.2 is deter-

mined by the RTD current voltage relationship

approximated by , where an external bias cur-

rent and V1,V2 > 0 have been assumed.

The basic idea of the circuit in Fig. 2 is roughly

the following. An input current source charges

the capacitance C up to a threshold value given

by J(V) of the RTD. A switching up occurs which,

however, will be inverted due to the LG time

constant. As a result, a spike is produced and

after an RC time constant another switching

occurs. Hence the spiking period is determined

by the amplitude of the input current. The trans-

C

R

GV

J(V) L

Fig. 2: Equivalent circuit of a

monostable RTD-NLTL

mission line itself ensures the generation and

propagation of identical pulses - such as solito-

ns - formed after a few diodes.

ResultsFig.3 shows a numerical result for an input

sinusoidal wave of 25 GHz. As can be seen, the

monostable RTD-NLTL produces a set of 6 puls-

es per period. When the input frequency or the

input amplitude are changed, the number,

phase, and position of the spikes are altered in

a characteristic way.

Fig.4 shows an example where the width and

amplitude of a rectangular input signal have been

changed. As a result, the generated spiking as

obvious from the regions with different shadings

is a characteristic pattern for the input signal. In

particular, we observe that the number of spikes

per time depends linearly on the applied current

amplitude providing a linear voltage-frequency

conversion.

Such a NLTL can be used to realise a high

speed n-bit A/D converter similar to the lumped

version in [7-9]. Correspondingly, we propose

10 20 30 40 50 60 70

-1,0

-0,5

0,0

0,5

1,0

Time

Fig. 3: Generation of five pulses

per period of sinusoidal input wave (dashed

line)

3 RESEARCH60

to realise a common n-channel (for n bits) copla-

nar signal devider to provide input amplitudes by

powers of 2. Hence each channel delivers a spike

train to the output array establishing a Gray code

as in Ref.[7]. In the present case, the LC time

constant will determine the bandwidth which ex-

ceeds 100 GHz in the device under test.

ConclusionIn conclusion, a novel monostable RTD-NLTL

in MMIC technology is proposed which can gen-

erate a characteristic pulse pattern for a given

input signal. The application of such a NLTL for

an ultrafast A/D conversion is discussed in a

second step. The use of RTDs in the presented

circuit is expected to yield a bandwidth in ex-

cess of 100 GHz.

References[1] E.R.Brown, J.R.Söderström, and T.C.McGill,

Oscillations up to 712 GHz in InAs/AlSb

Resonant-Tunneling Diodes, Appl.Phys.Lett.,

vol.58, no. 20, pp. 2291-2293, 1991

[2] R.Sun, O.Boric-Lubecke, D.-S.Pan, and T.Itoh,

Considerations and Simulations of

Subfrequency Excitation of Series Integrated

Resonant Tunneling Diodes Oscillator,

IEEE Trans. Microwave Theory Techn., vol. MTT

- 43, no.10, pp. 2478-2485, 1995

[3] I.V.Ryjenkova, V. K.Mezentsev, S.L.Musher,

S.K.Turitsyn, R.Hülsewede, and D.Jäger, Milli-


sion Lines, Proc.1996 International Workshop

on Millimeter Waves, April 11-12, Orvieto, Italy,

1996

[4] [4] I.V.Ryjenkova, V.K.Mezentsev, S.L.Musher,

S.K.Turitsyn, R.Hülsewede, and D.Jäger, Milli-


sion Lines, Ann. Telecomm., Special Issue,

vol.52, No 3-4, pp. 134-139, 1997

[5] H.D.Crane, Neuristor - A Novel Device and Sys-

tem Concept, Proc. IRE, vol.50, pp. 2048-

2060, 1962

[6] J.Nagumo, S.Arimoto, and S.Yoshizawa, An Ac-

tive Pulse Transmission Line Simulating Nerve

Axon, Proc. of the IRE, vol.50, p.2061, 1962

[7] T.-H..Kuo, H.C.Lin, R.C.Potter, D.Shupe, A

Novel A/D Converter Using Resonant Tunnel-

ing Diodes, IEEE Journal of Solid-State Circuits,

vol.26, No.2, pp.145-149, 1991

[8] S.-J.Wei, H.C.Lin, R.C.Potter, D.Shupe, Dy-

namic Hysteresis of the RTD Folding Circuit and

ist Limitation on the A/D Converter, IEEE Trans-

action on Circuits and Systems II: Analog and

Digital Signal Processing, vol.39, No.4, pp.247-

251, 1992

[9] S.-J.Wei, H.C.Lin, R.C.Potter, D.Shupe, A Self-

Latching A/D Converter Using Resonant Tunnel-

ing Diodes, IEEE Journal of Solid-State Circuits,

vol.28, No.6, pp.697-700, 1993


2.00 3.00 4.00 5.00 6.00 7.00 8.002.00

3.00

4.00

5.00

6.00

7.00

8.00

9.00

10.00

Fig. 4: Contour plot of generated

number of spikes (see text)

3.4 Optical Sensor Systems

3.4.1 MQW-Electroabsorption-Modu-lator for Application in a fiberopticfieldsensor

M. SCHMIDT, R. HEINZELMANN, AND A. STÖHR

n this report we present electroabsorp-

tion waveguide-modulators for an op-

eration wavelength of 1.55µm. These devices

are fabricated for an application in a fiberop-

tical E-field sensor system [1], [2]. In this

system the task of the modulator is to con-

vert electrical signals with frequencies up to

6 GHz into optical signals.

IntroductionIn recent years there has been an increasing

interest in electrooptical modulators. The main

application of these devices is in fiberoptical

communication systems for the external modu-

lation of laserdiodes. Electroptical modulators

have been realised in lithium niobate as well as

in semiconductors using the Franz-Keldish-ef-

fect in bulk materials and the quantum confined

starck effect in multiple quantum well structures.

As MQW structures exhibit the strongest elec-

trooptic effect they allow the use of smaller elec-

trodes than the other mentioned modulator prin-

ciples. The resulting lower capacitance has the

advantage of higher cut off frequencies. Further-

more MQW modulators can be made insensi-

tive to the polarisation of the modulated light by

introducing tensile strain to the quantum wells.

This avoids the need for expensive polarisation

maintaining fibers.

InGaAs/InAlAs - MQW

nid InAlAs

n InAlAs

s. i. InP

+

n InAlAs-

nid InAlAs

contactLayer structure:

Fig. 1: Sketch of the electroabsorption waveguide modulator

3 RESEARCH62

I

Device structureIn Fig. 1 a sketch of the device is shown. The

modulators are grown by MBE in the ternary

material system InGaAs/InAlAs on InP subtrates.

The device structure is n+in -. The ridge

waveguide is formed by wet chemical etching

with cytric acid down to the n+ layer. Afterwards

the electrical contact are produced by vacuum

coating. The ohmic contact on the n+ layer is

realised in GeNiAu, for the Schottky contact on

the n- layer we use CrAu.

The design of the device structure was sup-

ported by BPM-Simulations and by calculations

of the electrooptical behaviour. For the BPM-

Simulations we used the comercial BPM-Soft-

ware BPM-Cad. The main aim of this simulation

was to determine the optical confinement fac-

tor, i. e. which part of the guided modes over-

laps with the absorbing MWQ layer. We calcu-

lated a confinement factor of 14 %. For the

calculation of the absorption coefficient of the

MQW-material in dependence of the electrical

field we used a transfer matrix method. From

the obtained results we calculated the optical

absorption of the modulator in dependence of

the applied electric field as shown

in Fig. 2.

Experimental resultsThe epitaxial wafers were char-

acterized by photoluminescence

measurements. The point of in-

terest was the spectral position of

the excitonic peak of the MQW

material, which indicates the po-

sition of the absorption edge. A

comparison between the experi-

mental results and the calcula-

tions is shown in Fig. 3.

The optical transmission of the

modulator is characterized by

coupling the light of an erbium

doped fiber laser into the

waveguide and detecting the

transmitted light on the other fac-

0 %

20 %

40 %

60 %

80 %

100 %0 5 10 15

Field (10 6 V / m)

Abs

orpt

ion

0 2 4 6Voltage [V]

Slope: 0,38 / V

Wavelength: 1.55µmDevice lenght: 500 µm

Fig. 2: Calculated electrooptical behaviour

of a modulator device.

6 7 8 9 101340

1360

1380

1400

1420

1440

1460

1480

1500

1520

1540

1560

1580

Mod12Shg 08

Mod 08 Mod 13 Shg 03

Mod 07Mod 05 Shg 07

Shg 04Shg 05

Mod 09

Exc

itoni

c w

avel

engt

h [n

m]

Thickness of quantum-wells [nm]

PL-Measurement at 12 KSimulation at 12 KSimulation at 300 K

Shg 06

Fig. 3: Excitonic wavelength of the MQW material determined

by photoluminescence measurements compared with calculated

values.

3.4 Optical Sensor Systems 63

et of the waveguide. The optical tranmission is

measured in dependence of the applied voltage

as shown in Fig. 4.

ConclusionsAn electrooptical MQW waveguide modula-

tor has been designed. The epitaxial layers have

been grown by MBE and characterised by pho-

toluminescence measurements. The position of

the measured exciton peaks was in good agree-

ment with the calculated values. Modulator de-

vices have been processed from these wafers

by wet chemical etching and vacuum coating of

the electrical contacts. The devices have been

characterized by optical transmisson measure-

ments.

References[1] Stöhr, R. Heinzelmann, T. Alder, M. Schmidt, M.

Groß, and D. Jäger, Integrated Optical E-Field

Sensors using TW EA-Modulators,

Interational Topical Workshop on Contemporary

Photonic Technologies CPT98, Technical Di-

gest, Tokyo, January 1998

[2] Heinzelmann, A. Stöhr, M. Groß, D. Kalinowski,

T. Alder, M. Schmidt, and D. Jäger, Optically

Powered Remote Optical Field Sensor Sys-

tem using an Electroabsorption-Modulator,

IEEE MTT-S International Microwave Sympo-

sium, Conference Proceedings, Baltimore, June

1998

3.4.2 Photovoltaic cells for fiber opticEMC - Sensor power supply

D. KALINOWSKI

hotovoltaic cells play an important

role in power supply of hybrid sen-

sors. A photovoltaic cell array is under con-

struction to supply an active fiber optic hybrid

sensor head. A prototype with first

experimental results will be shown.

IntroductionDue to more and more restrictive laws regu-

lating the electromagnetic compatibility (EMC)

of electronic equipment the necessity of devel-

oping precise and reliable sensors to measure

electromagnetic fields steadily increases. One

request for such sensors is non invasiveness.

Hence, our approach to reach this goal is to

develop a hybrid fiber optic fiels sensor. This

concept takes advantage of the fact, that opti-

cal fibers do not interfere with the electromag-

netic field that is to measure, but that they are

capable to transmit optical information. By this

distortion of the E-field is minimized. The photo-

voltaic cell array (PVC) described in this article

is part of this optical E-field sensor which is

shown in Fig. 1.

0 -1 -2 -3 -40,0

0,1

0,2

0,3

0,4

0,5

Mod

ulat

ion

[a.u

.]

Voltage [V]

λ = 1550 nm

Fig. 4: Modulation of the MQW modulator

versus applied voltage.

3 RESEARCH64

P

DeviceOne requirement to be matched by the PVC

is a high efficient conversion of optical into elec-

trical power. Therefore, special effort has to be

laid upon the layout and the composition of the

heterostructures. Since cheap and powerful la-

ser diodes are available in the 800 to 850 nm

wavelength regime and since GaAs has its opti-

mum photovoltaic response at about 800 nm,

the PVCs are designed as AlGaAs/GaAs pin-

diodes. Fig. 2 shows a photograph of a cell ar-

ray consisting of 4 cells. The diameter of the

active region is 600

µm, i.e. the diameter of

the core of the multi-

mode fiber used. Thus,

each quarter of this

array, i.e. each PVC, is

illuminated uniformly

leading to a maximum

generation of electrical

power by this configu-

ration [1].

The GaAs and AlGaAs layers are MBE grown

on semi-insulating GaAs substrate. The layer

structure, illustrated in Fig. 3, consists of an 100

nm n+-AlGaAs contact layer, a 2 µm i-GaAs ab-

sorption layer and a 100 nm p+-AlGaAs window

layer. A 10 nm p+-GaAs contact layer offers an

aluminium protection from oxidation.

The metallic contacts act as ohmic contacts.

GeNiAu is used for the n-contact and PtTiPtAu

for the p-contact.

Fig. 1: Sketch of the optically powered integrated optical field sensor

Fig. 2: Perspective view of the photovoltaic

cell array consisting of 4 cells

Fig. 3: Cross section of the photovoltaic cells


ResultsThe cell array is illuminated by a laser with a

emission wavelength of 800 nm. Measurements

of the max. efficiency show results up to 28%

depending on the optical input power (Fig. 4).

ConclusionThis report presents a photovoltaic cell array

for power supply of our hybrid EMC - Sensor.

The design and a first result are shown.

References[1] M. B. Spitzer, et. al., Monolithic series-con-

nected gallium arsenide converter development,

Proc. 22nd IEEE Photovoltaic Specialists Confer-

ence, Las Vegas, USA, 1991

3.4.3 Time- and frequency-domainelectro-optic field mapping of nonlin-ear transmission lines

P. BUSSEK, TH. BRAASCH, AND G. DAVID

e report on the measurements of

electric field distributions in mono-

lithic microwave integrated circuits (MMICs)

using electro-optic probing techniques. In

1996 the activities have been focused on the

analysis of microwave propagation in non-

linear transmission lines (NLTLs). The mea-

surements have been performed in frequency

domain as well as in time domain as they

have been done in one dimension as well as

in two dimensions. As an example, in this

documentation we present experimental

results of periodic NLTLs demonstrating the

generation of higher harmonics on these

devices and the formation of shock waves.

IntroductionIn recent years, the complexity of monolithic

microwave integrated circuits (MMICs) expand-

ed necessitating the development of measure-

ment techniques which keep abreast of the in-

creased demands of an appropriate

characterization of these devices. So far, net-

work-analyzers (NWA) are mostly used for on-

wafer microwave characterization of MMICs.

This measurement technique is well established

but its application is limited due to the fact that

the on-wafer probes needed for this technique

only allow the access to external ports. Thus,

no circuit-internal measurement or local failure

test of the device nore the observation of wave

propagation effects is possible using NWAs.

In contrast, electro-optic sampling has be-

come a sophisticated technique to study quan-

Fig. 4: Photovoltaic cell efficiency

3 RESEARCH66

W

titatively field distributions and wave propagation

effects insight microwave and millimeter-wave

devices [1-3]. This technique can be performed

in frequency as in time domain enabling the

detection of the amplitude and phase of a

microwave signal as of the temporal evolution

of this signal. The spatial resolution of this meth-

od is measured to be down to less than 0.5 µm

[4]. Hence, each MMIC component can be test-

ed and evaluated noninvasively up to millime-

ter-wave frequencies. By combining the direct

electro-optic probing with 2D scanning of the

laser beam two-dimensional field mappings of

the device under test (DUT) are possible [5].

Thus, wave propagation effects can be studied

which gain a growing interest by circuit design-

ers for two reasons. On one hand, these effects

influence the electrical behaviour of MMICs re-

sulting in a limitation of their bandwidth or the

generation of unwanted modes [3]. On the other

hand, novel types of integrated circuits such as

nonlinear transmission lines (NLTLs) can be

designed which make use of these effects, e.g.

to generate short electrical pulses or to excite

higher harmonics. This has been observed par-

ticularly in periodic NLTLs and will be shown in

this report.

Experimental setupThe experimental setups used in this project

are sketched in Fig. 1. An actively modelocked

Nd:YAG laser (wavelength = 1064 nm, pulse

repetition rate = 82 MHz) combined with a fiber-

grating pulse compressor provides short pulses

of 5 ps FWHM (full width at half maximum) cor-

responding to a bandwidth of the setup in ex-

cess of 80 GHz. The device under test (DUT) is

illuminated from the backside, i.e. the direct elec-

Fig. 1: Experimental setup, (a) for the frequency domain measurements, (b) for the time domain

measurements


tro-optic sampling is applied since the

linear electro-optic effect in the sub-

strate itself is used for the modulation

of the polarization. To convert this po-

larization modulation into an intensity

modulation, polarizers and a quarter-

wave plate are implemented in the op-

tical pathway. The reflected intensity is

detected by a small area photodiode.

Due to the combination of a small area

photodiode and the confocal arrange-

ment of the setup out-of-focus-light is

suppressed improving the spatial res-

olution of the measurement system

down to less than 0.5 µm [4]. For 2D

scans the probe stage is movable in

the x- and y-direction.

Fig 1(a) illustrates the experimental setup for

frequency domain measurements. Here, a spec-

trum analyzer working as a tunable bandpass is

used for the detection of the signal amplitude.

The intermediate frequency is set to several

MHz, since in this regime the high speed ava-

lanche photodiode used exhibits a maximum

sensitivity. The microwave synthesizer, the mod-

elocker synthesizer of the laser system and the

spectrum analyzer are phase stabilized via a

phase locked loop (PLL). For the phase mea-

surements the spectrum analyzer is replaced by

a lock-in amplifier (not shown in Fig. 1(a)). In a

second mode this setup is used to receive an

optical image of the measurement region by sim-

ply detecting its front surface reflectivity. Thus,

the electro-optical signal can be normalized to

the particular reflectivity of the device, and the

absolute value of the voltage between the de-

vices top and bottom surface can be determined

[6].

For the measurements performed in time

domain some modifications of the experimental

setup are needed. As depicted in Fig. 1(b), the

optical pulses themselves generate the electri-

cal signal in order to establish a phase locking

between the probe pulses and the electrical mi-

crowave signal. A small part of the output beam

of the Nd:YAG laser is separated via a beam

splitter and chopped at about 4 kHz. The photo-

current of a fast photodiode detecting this out-

put signal then traverses a mechanical delay line

that periodically shifts the phase of the signal

while the observation point is kept constant. The

photodiode has to be changed by a slow Ge-

diode to apply lock-in techniques at the chop-

ping frequency.

Frequency domain measurementsThe results presented here all have been

done with periodic nonlinear transmission lines.

For a more detailed description of the examined

samples see R. Hülsewede, Investigations of

0 500-140

-130

-120

-110

-100

-90

-80

1500 2500propagation distance (µm)

15 GHz

30 GHz

45 GHz60 GHz

Fig. 2: Electro-optic signal of the fundamental microwave

at 15 GHz and of its higher harmonics along the center

conductor of a periodic NLTL from input to output.

3 RESEARCH68

pulse compression on nonlinear transmission

lines, in this annual report. Fig. 2 depicts the

spatial distribution of the incident fundamental

electrical signal at 15 GHz and the amplitudes

of the second, third, and fourth harmonic with

frequencies up to 60 GHz. As can be seen the

amplitude of the fundamental signal decreases

in the direction of propagation whereas the am-

plitudes of the

higher harmon-

ics increase in-

dicating that

they are

g e n e r a t e d

along the trans-

mission line.

The obvious

standing wave

patterns are

caused by an

i m p e d a n c e

mismatch at

the end of the

line and phase

mismatching of

the harmonics.

Two-d imen-

sional field

mappings of

an NLTL are

shown in figs.

3. Here, the

frequency of

the fundamen-

tal is 6 GHz,

and the metal-

lization struc-

ture of the de-

vice, the electro-optic signal of the fundamental,

the second harmonic at 12 GHz and the third

harmonic at 18 GHz are presented in Figs. 3(a)

- (d), respectively. These Figs. show the decrease

of the fundamental signal and the increase of the

harmonics while propagating along the NLTL as

Fig. 2 does, but additionally they reveal an un-

symmetrical distribution of the electro-optic signal

Fig. 3: Nonlinear transmission line; (a) metallization structure; results of 2D field

mappings (b) at the fundamental at 6 GHz, (c) at the second harmonuc at 12 GHz

and (d) at the third harmonic at 18 GHz.


that can not be detected with one-dimensional

linescans as is the case in Fig. 2. We contribute

this behaviour to the excitation of parasitic prop-

agation modes [3].

Time domain measurementsIn time domain measurements there is a fixed

phase relation between each particular mea-

surement point. Thus, the evolution of a period-

ic signal can be observed as is elucidated in

Figs. 4(a) to 4(d) for the development of a sinu-

soidal electrical signal of 6 GHz propagating

along a periodic NLTL at the 1st, the 5th, the

10th and the 14th diode, respectively. As can

be seen, shock waves are generated with fall

times down to 5 ps due to the interaction of the

higher harmonics. In the

time domain, this forma-

tion of a shock wave is the

counterpart to the

generation of harmonics in

the frequency domain.

The presented results

validate that the electro-

optic probing technique is

capable of studying and

demonstrating this effect

as well.

ConclusionIn summary, electro-

optic measurement tech-

niques have been used

to internally investigate

wave propagation effects

along periodic nonlinear

transmission lines en-

abling circuit-designers to

get an insight into the in-

circuit electrical characteristics of complex mi-

crowave devices. The generation of harmonics

and the formation of shock waves have been

demonstrated showing, that this method is suit-

able to examine internal field distributions in

MMICs in both, frequency domain and time do-

main.

References[1] K.J. Weingarten, M.J.W. Rodwell, and D. Bloom,

Picosecond optical sampling of GaAs integrated

circuits, IEEE J. Quantum Electron., vol. QE-

24, (1988), pp. 198-220

[2] G. David, S. Redlich, W. Mertin, R.M. Bertenburg,

S. Kosslowski, F.J. Tegude, E. Kubalek, and D.

Jäger (1993), Two-dimensional direct electro-

-80 -40 0 40 80

k = 1

(a)

-80 -40 0 40 80

k = 5

(b)

-80 -40 0 40 80

k = 10

(c)

-40 0 40 80-80

k = 14

(d)

time (ps) time (ps)

time (ps) time (ps)

Fig. 4: Electro-optic signal of the fundamental microwave at 15 GHz and

of its higher harmonics along the center conductor of a periodic NLTL

from input to output.

3 RESEARCH70

optic field mapping in a monolithic integrated

GaAs amplifier, Proc. 23rd EuMC 1993, Madrid,

Spain, 1993, pp. 497-499

[3] G. David, R. Tempel, I. Wolff, and D. Jäger,

Analysis of microwave propagation effects us-

ing 2D electro-optic field mapping techniques,

Optical and Quantum Electronics, Special Issue

on Optical Probing of Ultrafast Devices and In-

tegrated Circuits, vol. 28, 1996, pp. 919-931

[4] G. David, P. Bussek, U. Auer, F.J. Tegude, and

D. Jäger, Electro-optic probing of RF signals in

submicrometre MMIC devices, Electron. Lett.,

1995, Vol. 31, No. 25, pp. 2188-2189

[5] M.G. Li, E.A. Chauchard, C.H. Lee, and H.-L.A.

Hung, Two-dimensional field mapping of GaAs

microstrip circuit by electrooptic sensing, Proc.

OSA Int. Top. Meeting `Picosecond Electronics

and Optoelectronics`, March 13-15, 1991, Salt

Lake City, USA, pp. 54-58

[6] G. David, W. Schröder, D. Jäger, and I. Wolff,

2D electro-optic probing combined with field

theory based multimode wave amplitude extrac-

tion: a new approach to on-wafer measurement,

Symposium Digest 1995 IEEE MTT-S Interna-

tional Symposium, May 15 -19, 1995, Orlando,

USA, pp. 1049-1052

3.4.4 Characterization of monolithicmicrowave integrated circuits byheterodyne electro-optic sampling

TH. BRAASCH

he propagation of electric signals in

the millimeter- and microwave regime

along monolithic microwave integrated cir-

cuits (MMICs) can be studied by electro-op-

tic measurement techniques. In this paper,

we describe the implementation of a hetero-

dyne electro-optic measurement setup and

present first experimental results.

IntroductionUsing common network analyzer methodes

(NWA) for the characterization of MMICs, the

device under test (DUT) is measured as an in-

tegral device and no in-circuit measurements are

possible [1]. The increasing working frequencies

of integrated circuits up to the millimeter- and

microwave region [2-4] necessitate measure-

ment techniques that allow an insight into the

device. In recent years, electro-optic sampling

(EOS) has become a sophisticated technique

to observe field distributions in MMICs with a

spatial resolution down to less than 0.5µm [5,

6]. Quantitative characterizations have been

carried out, and one- as two-dimensional mea-

surements are possible in time-domain as in fre-

quency domain [7, 8]. Thus, circuit-designers get

a knowledge of circuit-internal parameters, which

is of increasing importance since the complexi-

ty of the devices expands. Nevertheless, so far

the EOS has been mostly performed with a

pulsed laser. Here, to convert the microwave

down to frequencies, where spectrum analyzer,

lock-in amplifier and the photodiode used are

able to detect the electro-optic signal, the n-th

harmonic of the repetition frequency of the

pulsed laser is used to interact with the electric

signal. The electric bandwidth of these setups

is limited by the pulse width of the laser pulses.

In frequency domain, measurements on nonlin-

ear transmission lines up to 100 GHz are re-

ported [9, 10], and fall times down to 1.5 ps have

been measured in time domain [11]. However,

utilizing the n-th harmonic of the repetition fre-

quency of the pulsed laser for the down conver-

sion of the electric signal leads to a reduction of

the signal to noise ratio of the electro-optic sig-


T

nal since the phase noise of the setup increases

with the measurement frequency. To circumvent

this restriction two cw lasers can be used where

the second laser acts as a local oscillator. Here,

phase noise of the setup only depends on the

stability of the two lasers [12]. In this paper, we

present our measurement setup and show first

experimental results.

Experimental setupWe operate with two identical Er-doped fiber

lasers exhibiting a linewidth < 10 kHz. Both la-

sers are continuously tunable between 1530 nm

and 1560 nm leading to beat frequencies up to

4 THz. They deliver > 20 mW optical output and

show almost no mode-hops once they reached

thermal equilibrium. The degree of polarization

is > 99%. Fig. 1 demonstrates the configuration

of our setup. A small percentage of both lasers

is coupled into a Fabry-Perot optical spectrum

analyzer with a finesse >150 for the detection

of the beat fre-

quency between

the two lasers.

After passing a l/

4 polarization

control the light of

the first laser is

back-side cou-

pled into the de-

vice under test

where the stray

field of the micro-

wave in the sub-

strate interacts

with the laser

light via the Pock-

els-effect. The

reflected light is

again coupled

into the fiber and traverses the circulator and a

second polarization control. Thus, the polariza-

tion modulation due to the Pockels-effect is

converted into an intensity modulation directly cor-

related to the strength of the electric field at the

particular point of measurement. The DUT is

placed on a translational stage enabling two-

dimensional field mappings of the field distribu-

tion. Due to the confocal arrangement of the setup

it can also be used as an optical microscope. In

this mode, the front surface reflectivity of the

device can be detected and afterwards the elec-

tro-optic signal can be normalized to the particu-

lar reflectivity at each particular measuring point.

As a consequence, the absolute value of the

voltage between the devices top and bottom side

can be determined [8]. Via a fiber coupler the

local oscillator, i.e. the second laser, is

superposed to the reflected light from the DUT

carrying the information of the microwave signal

applied to the DUT. The intermediate frequency

Fig. 1: Sketch of the heterodyne electro-optic measurement setup

3 RESEARCH72

between the second laser and one sideband of

the first laser, i.e. f1 ± fm with f1 the frequency of

the first laser and fm the microwave frequency,

can now be adjusted by the frequency f2 of the

second laser. This intermediate frequency is de-

tected by a fast travelling-wave photodetector

[3]. Hence, any microwave frequency within the

tuning range of the two lasers can be converted

to some MHz or GHz only affected by the inherent

phase noise of the two lasers but independently

of the frequency.

ResultsA coplanar waveguide structure (CPW) was

used to demonstrate the feasibility of the setup

as a scanning microscope. Fig. 2 depicts the

surface reflectivity of the CPW. The difference

frequency between the lasers was arbitrarily set

to 8.5 GHz since at this value they worked ex-

tremely stable and the detected signal of the

spectrum analyzer was about 50 dB larger than

the noise floor. In the next step, a microwave

has now to be applied to the device and the elec-

tro-optic signal has to be detected as in [6], [8]

or [10] with the pulsed laser system.

ConclusionIn summary, to bypass the phase noise re-

strictions of a pulsed electro-optic measurement

setup two narrow linewidth tunable cw lasers

have been implemented in the configuration.

Thus, the phase noise only depends on the char-

acteristics of the lasers but is not affected by

the measurement frequency. Owing to the tun-

ing range > 30 nm of the Er-doped fieber lasers

used heterodyne detections of electric signals

up to 4 THz should be possible. As a first result,

the surface reflectivity of a coplanar waveguide

structure detected at 8.5 GHz difference fre-

quency is presented.

References[1] D.J. Bannister and M. Perkins, Tracebility for on-

wafer s-parameter measurements, IEE Proc. A,

vol. 139, 5, 1992, pp. 232-233

Fig. 2: (a) Sketch of a coplanar waveguide structure, (b) reflected intensity measured with the hetero-

dyne setup at 8.5 GHz.


[2] M.J.W. Rodwell, S.T. Allen, R.Y. Yu, M.G. Case,

U. Bhattacharya, M. Reddy, E. Carman, M.

Kamegawa, Y. Konishi, J. Pusl, and R. Pullela,

Active and nonlinear wave propagation devices

in ultrafast electronics and optoelectronics, Proc.

IEEE, vol. 82, 7, 1994, pp. 1037-1060

[3] M. Alles, Th. Braasch, R. Heinzelmann, A. Stöhr,

and D. Jäger, Optoelectronic devices for micro-

wave and millimeterwave optical links, Proc.

MIKON96, Workshop Optoelectronics in Micro-

wave Technology, Warsaw, Poland, 1996 (in-

vited)

[4] I.V. Ryjenkova, V.K. Mezentsev, S.L. Musher,

S.K. Turitsyn, R. Hülsewede, and D. Jäger, Mil-

limeter wave generation on nonlinear transmis-

sion lines, Ann. des Telecomm., Special Issue,

1996 (invited)

[5] K.J. Weingarten, M.J.W. Rodwell, and D. Bloom,

Picosecond optical sampling of GaAs integrated

circuits, IEEE J. Quantum Electron., 1988, QE-

24, pp. 198-220

[6] G. David, P. Bussek, U. Auer, F.J. Tegude, and

D. Jäger, Electro-optic probing of RF signals in

submicrometre MMIC devices, Electron. Lett.,

1995, Vol. 31, No. 25, pp. 2188-2189

[7] M.G. Li, E.A. Chauchard, C.H. Lee, and H.-L.A.

Hung, Two-dimensional field mapping of GaAs

microstrip circuit by electrooptic sensing, OSA

Proc. Picosecond Electronics and Optoelectron-

ics, March 13-15, 1991, Salt Lake City, USA, pp.

54-58


Analysis of microwave propagation effects us-

ing 2D electro-optic field mapping techniques,

Optical and Quantum Electronics, Special Issue

on Optical Probing of Ultrafast Devices and In-

tegrated Circuits, 1996, 919-931

[9] R. Majidi-Ahy, B.A. Auld, and D.M. Bloom, 100

GHz on-wafer s-parameter measurements by

electro-optic sampling, IEEE MTT-S, 1989, pp.

299-302

[10] Th. Braasch, G. David, R. Hülsewede, U. Auer,

F.-J. Tegude, and D. Jäger, Propagation of mi-

crowaves in MMICs studied by time- and fre-

quency-domain electro-optic field mapping, Proc.

Trends in Optics and Photonics Series (TOPS)

of OSA 1997 Spring Topical Meeting Ultrafast

Electronics and Optoelectronics, 1997, Lake

Tahoe, USA

[11] K.S. Giboney, S.T. Allen, M.J.W. Rodwell, and

J.E. Bowers, Picosecond measurements by free-

running electro-optic sampling, Phot. Tech. Lett.,

vol. 6, 11, 1994, pp. 1353-1355

[12] S. Loualiche and F. Clerot, Electro-optic micro-

wave measurements in the frequency domain,

Appl. Phys. Lett. 61, (18), 1992, pp. 2153-2155

3.4.5 Development of an experimen-tal setup for field probe measure-ments on nonlinear transmissionlines

D. KALINOWSKI AND R. HÜLSEWEDE

n experimental setup for field probe

measurements has been established.

The electrical fields on nonlinear transmis-

sion lines has been measured up to 60GHz.

Theoretical considerations have been done

up to 200GHz. Experimental results have

been compared with results using electro-

optic testing.

IntroductionIn recent years there has been a great

progress in the development on nonlinear trans-

3 RESEARCH74

A

RsRv jXAUUl

Fig. 2: Equivalent circuit for the electric field

probe

mission lines making them to key structures for

future microwave circuits [1]. To characterize

these structures measurements at external ports

are not sufficient. Different noncontacting probes

give a chance to take a look at the field distribu-

tion on the transmission lines. The probes are

the electro-optic probe [2], the magnetic field

probe [3] and the electric field probe [4]. Such

an electric field probe has been established. Its

potential has been demonstrated.

Experimental setupA sketch of the experimental set-up is shown

in Fig. 1. The nonlinear transmission lines

(NLTLs) are supplied by a microwave syn-

thesizer and a DC-voltage source. They can be

loaded with variable resistance. The probe de-

tects the electric field above the NLTL. Its sig-

nal is evaluated by a spectrum analyzer. A com-

puter controls the position of the probe and

stores the data measured by the spectrum ana-

lyzer. By that way two-dimensional field map-

pings can be done.

Theoretical resultsThe probe can be described by an equiva-

lent circuit which is shown in Fig.2 [5]. The volt-

age Ul depends on the electric field strength.

The probe impedance is given by RV and RS which

characterize the thermal and radiation losses and

XA describing the open line.

The relation between the indicated power at

the spectrum analyser and the square of the

measured field intensitity is shown in Fig. 3. It

demonstrates the steady increasing probe sen-

sitivity versus frequency. Thus the probe can

be used over the whole frequency range.

Measurements confirm this equivalent circuit.

Fig. 4 shows the relation between the indicated

signal on the spectrum analyser and the signal

frequency. The experimental results are given

by the dots (·), the theoretical results by the line

(-). A good correspondence is given between

these results. So the equivalent circuit can be

used to determine the electric field strength by

the spectrum analyzer signal.

0 40 80 120 160 2000

0,2

0,4

0,6

0,8

frequency (GHz)

sens

itiv

ity

(

)

pA Vm

Fig. 3: Sensitivity versus frequency.


xy

z

NLTL

mixer synthesizer

dc - source

load

spectrum-analyser

bias tee

prober

Fig. 1: Sketch of the experimental Setup

Fig. 5:2-dim. field mapping of an NLTL (a)

signal with 7.4 GHz, (b) generated 3rd

harmonic at 22.2 GHz c) Sketch of the NLTL

Experimental resultsOne experimental result achieved with this

setup is presented in Fig. 5. A 4 µm NLTL has

been examined. The sketch of this line is shown

in Fig. 5c. Whereas one end has been connect-

ed with a synthesizer, the other end has been

unloaded. The synthesizer has supplied the line

with a 7.4 GHz microwave signal. The field dis-

tribution at this frequency is shown in Fig. 5a.

The 3rd harmonic generated on the line is shown

in Fig. 5b. The amplitude increases in the direc-

tion of propagation indicating the generation

along the transmission line. Furthermore an

asymmetrical transversal distribution is revealed.

ConclusionMeasurements up to 60 GHz have been done

successfully. Field distributions on transmission

lines with electrode widths down to 12 µm has

been shown. Comparisons with theoretical re-

sults and electo-optic probing are in good agree-

ment.

References[1] M.J.W. Rodwell, et al., Active and nonlinear

wave propagation devices in ultrafast electron-

ics and optoelectronics, IEEE Proc., Vol. 82, No.

7, pp. 1037-1059, 1994

[2] P.Bussek, G. David, Quantitative analysis of

two-dimensional electro-optically measured field

distributions in MMIC-structures, Annual Report,

Gerhard-Mercator-Universität - GH - Duisburg,

Fachgebiet Optoelektronik, 1995

[3] Y. Gao, I. Wolff, A new miniature magnetic field

probe for measuring three-dimensional fields in

planar high-frequency circuits, IEEE Transac-

tions on Microwave Theory and Techniques, Vol.

44, No. 6, June 1996, pp. 911-918

3 RESEARCH76

5 10 15 20 25 30 35 40-50

-46

-42

-38

-34

-30

-26

measurement

frequency (GHz)

theory

rel.

sign

al (

dB)

3.5 Technologies for Optoelectronic Components and Systems 77

3.5 Technologies for Opto-electronic Components andSystems

3.5.1 Development of a measurementsystem for the optical characteriza-tion of full-colour-LED-displays

M. WENNING, R. BUß, AND A. STÖHR

or the optical characterization of a full-

colour-LED-display, two measurement

systems have been developed. The first one

is to determine the spatial distribution of a

LED and the second one is to receive the

spectrum of a LED or a LED-pixel. The x, y, z

colour coordinates of the CIE chromaticity

diagram are evaluated from the spectrum. By

using the measurement system, an

optimization of a full-colour LED-display was

performed.

IntroductionAmong the various types of flat panel displays

(i.e., CRT, VFD, PDP, LCD, LED and EL), LED

displays are widely used as information boards

and as transportation terminal displays due to

their excellent reliability, service life and visibili-

ty.

Particularly as a result of the remarkable

progress made with high-brightness blue and

green LEDs, full-colour displays can now be

established for outdoors. Any colour can be pro-

duced using the three primary colours red, green

and blue. In this report, measurement systems

are developed to characterize a full-colour LED-

display. Furthermore, the LUMINO XTralux ML

4 C-Pixel and an optimized Pixel has been char-

acterized.

[4] D. Kalinowski Entwicklung eines Feldsonden-

meßplatzes zur zweidimensionalen Analyse

elektromagnetischer Feldverteilungen auf

nichtlinearen Leitungen, Diploma thesis,

Gerhard-Mercator-Universität Duisburg, 1996

[5] R. Geißler, et al., Taschenbuch der Hochfre-

quenztechnik Band 2: Komponenten, Springer-

Verlag, Berlin-Heidelberg, 1992

F

Measurement systemsThe spatial distribution of a LED is measured

with the setup shown in Fig. 1. The LED is fixed

on a LED-holder and is driven by a constant

current. By rotating the

swivel-arm in 1°- steps,

the data of the spatial

distribution is received.

The Fig. 2 shows

the setup to determine

the spectral distribu-

tion of a LED or LED-

pixel. After the spec-

tum is measured, the

x, y, z colour coordi-

nates are determined

[1,2].

The CIE (Commis-

sion Internationale de

lEclairage) diagram is

the standard colouri-

metric system. The x, y, z axis of this diagram

are based on three colour-matching functions,

each of which is related to the spectrum of red,

green and blue. A sequence of single-wavelength

computer

IEEE

Lock-in-amplifierIn Ref.

chopper

single-LED orLED-Pixel

light stop with aperture

lens hole

detector =lens + photodiode

monochromatorwith

stepper motor

Fig. 2: Measurement setup - spectrum

3 RESEARCH78

0,0 0,2 0,4 0,6 0,80,0

0,2

0,4

0,6

0,8

yellowish greengreen

red

blue

single-wavelenght colours

D65

y

x

Fig. 3: CIE diagram of all LEDs

0,0 0,2 0,4 0,6 0,80,0

0,2

0,4

0,6

0,8single-wavelenght colours

D65

red

green

blue

y

x

0,0 0,2 0,4 0,6 0,80,0

0,2

0,4

0,6

0,8single-wavelenght colours

D65

red

green

blue

y

x

Fig. 4: (a) CIE diagram of LUMINO Xtralux-pixel and (b) of the optimized pixel.

colours can be expressed as a curve in the x, y,

z space of the CIE diagram and the projection of

the curve on the x, y plane is a horseshoe-shaped

pattern (Fig. 3). Any colour can be expressed as

a point inside of this horseshoe-shaped curve.

Experimental resultsThe colour coordinates of all measured LEDs

are shown in Fig. 3 which are determined from

the spectrum of the LEDs. In Fig. 4 (a) every

colour in the triangle region, of which the verti-

ces indicate the three primary colours of the

LUMINO -Xtralux LEDs, can be radiated by ad-

justing the luminous intensity of each LED. As

0,0 0,2 0,4 0,6 0,80,0

0,2

0,4

0,6

0,8

80°

60°

40°20°

0°


D65

y

x

0,0 0,2 0,4 0,6 0,80,0

0,2

0,4

0,6

0,8

90°

80°60°

40°20°0°


D65

y

x

Fig. 5: (a) Colour-shift of LUMINO-Xtralux-pixel and (b) of the optimized pixel.


(a) (b)

(a) (b)

0 20 40 60 800,0

0,2

0,4

0,6

0,8

1,0

green

blue

red

viewing angle /°

rela

tive

Int

ensi

ty

0 20 40 60 800,0

0,2

0,4

0,6

0,8

1,0

green

red

blue

viewing angle /°

rela

tive

inte

nsi

tyFig. 6: (a) Spatial distribution of the LUMINO-Xtralux-pixel and (b) of the optimized pixel.

seen in the diagram, it is not possible to obtain

the standard white D65, which is outside the tri-

angle. The colour coordinates of an optimised

Pixel are shown in Fig. 4 (b). The LEDs are well-

chosen to obtain a greater triangle region. In

Fig. 5 the colour coordinate variation versus the

viewing angle of these two Pixels is shown. The

XTralux ML 4 C-Pixel (Fig. 5 (a)) has a large co-

lour shift to the red primary colour, which can

explained with a wider spatial distribution of the

red LED (cf. Fig. 6 (a)).

The small shift of the optimized Pixel was

performed by using a red, green and blue LED

having nearly the same spatial distribution. This

was obtained by modification of the lense-form

(encapsulation) of the LED. Furthermore, the

LED-surface is roughend.

ConclusionsWithin the scope of this thesis, two measure-

ment setups have been developed. Further-

more, LEDs and the LUMINO XTralux ML 4 C-

Pixel have been characterized. With the

knowledge of colour metrics and the colour co-

ordinates of the LEDs, an optimized Pixel has

been assembled. A greater colour range and a

nearly constant colour coordinate versus view-

ing angle of the pixel has been achieved.

References[1] Heinweg Lang, Farbmetrik und Fernsehen,

R.Oldenburg München Wien, ISBN 3-486-

20661-3, 1977

[2] DIN 5033, Farbmessung

3.5.2 Opinion poll on the evaluationof the legibility of LED-based dis-plays

R. HEDTKE AND R. BUß

n this report the legibility of LED-based dis-

plays is evaluated on the base of

interviews with passengers of the public

local traffic. To reach this aim a model ex-

plaining the causal connection between

several external influences and the legibility

is made up according to DIN 1450. Based on

this model a questionnaire for the interviews

3 RESEARCH80

(a) (b)

I

is developed. The gained data of the inter-

views is evaluated using statistic methods.

IntroductionToday, the permanent availability of informa-

tion has become of increasing importance,

where the transmission of visual information cer-

tainly becomes to play a more and more impor-

tant role. Due to the permanent development in

the area of LED-technology it has become pos-

sible to produce so-called super-luminescence

light emitting diodes (SLED), having a very high

brightness. In the public local traffic LED-based

displays find an increasing application. In co-

operation with the company LUMINO/Krefeld, a

producer of such displays, methods to improve

the legibility of those display-sys-

tems (cf. Fig. 1) are acquired.

The causal connectionThe causal connection based

on DIN 1450 is modified referring

to the requirements of the ques-

tionnaire. The result is shown in

Fig. 2 where the arrows describe

the influences. It can be seen that

there are eight major points hav-

ing an influence on the legibility:

The type face and size, brightness and colour,

the distance between each letter, the distance

of view, the light conditions, and personal influ-

ences.

The used questionnaireBased on the causal connection, the ques-

tionnaire shown in Fig. 3 has been developed

to proof and determine the level of influence of

each point. The first three questions are so called

icebreaker-questions to start the conversation

with the person to be interviewed. The following

questions are related to the legibility of the dis-

played text. Legibility, type size, distance be-

tween each letter, type face, brightness, and

colour are assessed using marks between one

and five. This type of assessment has been cho-

sen because everyone

knows this marks from

school. This causes

that comprehension

problems are avoided.

Furthermore, tenden-

cies like small, right,

and large where re-

corded if possible. Fi-

nally, the other points

Fig. 1: LED-based display used in the public local traffic.

TYPE FACE BRIGHTNESS COLOUR

TYPE SIZE LEGIBILITYDISTANCE BETWEEN LETTERS

DISTANCE OF VIEW LIGHT PERSONALINFLUENCES

Fig. 2: Model of the causal connection.


Fig. 3: The used questionaire

3 RESEARCH82

of influence are recorded after the main part of

the interview.

InterviewFor the interviews, locations in the following

cities are selected according to the practicabili-

ty of the interviews and considering the local

conditions: Duisburg, Essen, Düsseldorf; Ober-

hausen, Leipzig, Stuttgart.

At each of these locations interviews with 50

passengers have been carried out. To reach a

high comparability of the gathered data, the in-

terviews only took place on platforms for the pub-

lic local traffic.

ConclusionThe Fachgebiet Optoelektronik at the Ger-

hard-Mercator-Universität Duisburg has inves-

tigated the judgement of the legibility of

LED-based displays in co-operation with the

company LUMINO/Krefeld, a producer of such

displays. The aim of this study was the record-

ing of subjective opinions of the users of the

public local traffic.

The legibility of the analysed LED-based dis-

plays has been valued by up to 90% of the in-

terviewed passengers with well or very well.

Especially the displays based on yellow LEDs

in the city of Leipzig have been rated very pos-

itively. The majority of the passengers have felt

the brightness to be right. As well in Leipzig, the

valuation has been the same, even if the dis-

play was exposed to direct sunlight. As a rule,

type size, letter distance, and type were also

valued as right. Up to 80% of the passengers

have felt as being informed very well by those

displays.

In summary, the users of the dynamic pas-

senger-information-system are contented with

the quality and information provided by these

systems. It should be pointed out that the LED-

technology serves the high requirements refer-

ring to the demands in the public local traffic.

Even under unfavourable conditions like direct

sunshine the judgement is good or very good.

References[1] DIN 1450, Beuth, Berlin, Juli 1993

[2] V. Dreier, Datenanalyse für Sozialwissen-

schaftler, Oldenbourg, München-Wien, 1994

[3] K. Holm (Hrsg.), Die Befragung 1, Franke, Mün-

chen, 1975

[4] E. Noelle, Umfragen in der Massengesellschaft,

Rowohlt, Reinbek bei Hamburg, 1963

3.5.3 Evaluation of possible improve-ments to enhance the UV-power effi-ciency of a xenon flashlamp system

B. NEUHAUS AND A. STÖHR

his paper will present experimental

results of the characterisation of a

xenon flashlamp system with a flat alumini-

um rear reflector fabricated by Bläsing Elek-

tronik GmbH. The spatial distribution of the

UV-radiation is determined. Studies about

typical gas components and some materials

of the discharge tube result in possibilities

to optimize the arc lamp. Furthermore the

shape of a single flash is recorded and the

efficiency is measured in dependence of the

pulse repetition rate. To optimize this sys-

tem several rear reflectors with different geo-

metrics are tested and the reflection index

of five different materials is measured in the

UV and NIR wavelength range. For all these

examinations different measurment setups

are worked out.


T

IntroductionThe UV-drying process is of great significance

to the industry working with printing procedures

e.g. the silk-screen printing. New UV-paints are

free from solvents and they harden only by irra-

diation with UV-light [1]. Therefore powerful

lamps with great emission in the UV-range are

very important. Pulsed UV-lamps offer several

advantages compared with conventional UV-

burners. With flashlamps the intense heat emis-

sion can be reduced; they radiate only for a short

pulse duration but the hardening process is more

effective because the radiation output of

flashlamps is greater than of conventional lamps

[2]. The UV-flash drying process will become

an economical and nonpolluting alternative pro-

cess in the future.

The intention of the following investigations

is to characterize such a flashlamp system and

to optimize it based on the experimental results.

Arc lamp construction and priniciple of aflashlamp system

Fig.1 shows the construction of a typical arc

lamp. The electrodes are set in a clear quartz

tube. The type of quartz depends on the desired

output spectrum. The electrodes are made of

tungsten to enhance

electron emission. Arc

lamps are filled with inert

gas under several atmo-

spheric pressure or a

mixture of gas and a def-

inite amount of mercury.

The internal pressure in

the tube increases during

operation to 15-75 bar,

depending on the lamp

type.

The flashlamp system operates by sending

an electric charge from a pulse generator to the

gas filled lamp. The gas absorbs the energy by

storing it in its atoms and subsequently it re-

leases the energy by emitting photons, which

results in a high intensity flash of light [3]. Light

emssions in all directions can be guided by in-

stalling a reflector behind the lamp which col-

lects the light and reflects it onto a surface which

should be treated. Caution: these lamps produce

high intensity UV radiation and ozon. Precau-

tions are necessary during operation mode.

Experimental setupsFig. 2 and Fig. 3 show the two basic experi-

mental setups used for the measurements. In

order to determine the spatial distribution of

emitted radiation and to determine the UV-effi-

ciency of the flashlamp in the UV-range the set-

up in Fig.2 is used. The pulse generator sup-

plies the flashlamp with electrical impulses. The

pulse repetion rate (1,56Hz, 3,12Hz, 6,25Hz,

12,5 Hz, 25 Hz) can be adjusted by a rotary

switch. The light pulses, radiated by the

3 RESEARCH84

cathode

anode

tube

filling gas

+

-

Fig. 1: Con-

struction of arc

lamps [2].

Fig. 2: Experimental setup to measure the

relative optical output of an arc flashlamp.

flashlamp, are detected by a special

silicon carbide (SiC; spectral range: 210-

380nm) photodiode.

Fig. 3 shows the setup to determine

the UV-reflectivity of different reflector

materials. The light beam of a mercury

lamp is deflected by a beam splitter in

definite directions. The reflector reflects

the incoming light beam which is then

focused on a SiC photodiode. By us-

ing the second detector it is possible

to compare the input power with the re-

flected power in order to determine the

reflection index.

For all investigations in the NIR

wavelength range a monochromator

and a silicon photodiode (BPW20; 375-1100nm)

are used to receive spectral values; the light

source is a halogene lamp. To record a single

pulse an oscilloscope is used.

Experimental resultsa) Fig. 4 shows the relative optical output pow-

er with respect to time of the flashlamp type

UVL-1500/TP1 fabricated by Bläsing Elektron-

ik GmbH. The pulse shape has a fast rise time

t r( % %)10 90− of 31,5ms and slower decay. The

pulsewidth t FWHM is 520ms. These are typical

values for flashlamps [2]. Further experiments

have shown, that the flash energy as well as

the pulswidth stay constant for all frequencies

in the range from 1,56Hz until 25Hz.

b) In order to determine the spatial distribution

of the radiation from the flashlamp type UVL-

1500/TP1 the SiC-photodiode is led along a

semicircle around the lamp as shown in Fig.

5. One can record the radiant intensity (rela-

tive) for a number of angles. The radius is r =

100cm. The values are recorded in steps of

one degree and they are related to the 90°-

direction. Fig. 6 shows the results (normal

curve) given in a polar coordinate system. It

can be seen that the distribition of the radia-

tion is symmetrical to the axes. Besides, it

can be noticed that over an angle of a= 168°

fifty percent of the radiant power compared

with the power in direction of 90° is still radiat-


Fig. 3: Experimental setup to measure the

reflection index.

t sFWHM = 520µ

t sr( ,10% 90%) 315− = µ

− 2 5. ms 2 5. ms0 0. s

500µs div/

op

tica

l po

wer

(r

el.)

0

t ime t

Fig. 4: Relative optical output power of the flashlamp

UVL-1500/TP1.

ed by the flashlamp. So the angle of radiation

is very wide. Mostly such a wide angle is un-

desirable because one looses most of the

radiant in the borderlands when no object is

placed very closed to the radiant source. For

plenty of applications a distribution like a club

(dashed curve) is desired. It is possible to

achieve such a distribution with special rear

reflectors.

c) Fig. 7 represents the UV-efficiency in addic-

tion to the pulse repetition rate from the

flashlamp type UVL-1500/TP1. The values

are calculated for a wavelength range from

210-380nm as well as for the range from 270-

280nm. The detector was about two meters

away from the light source and the measure-

ments were repeated several times for all rep-

etition rates which are adjustable. From these

values a mean value is formed for each fre-

quency. This mean value is used for the cal-

culation of the efficiency for flashlamps. The

illustration shows that the efficiency stays

nearly constant for each frequency. The con-

3 RESEARCH86

flashlampUVL 1500-TP/1

SiC-detectorJEC-1210-380nm

r = 100cm

Fig. 5: Principle to record the spatial distribu-

tion of the radiant.

Fig. 6: Principle to record the spatial distri-

bution of the radiant.

effic

ienc

y

ηUV270 280−

ηUV210 380−

η ⋅ −( )10 9

pulse repetition rate f (Hz)Pu ls

Fig. 7: UV-efficiency in addiction to the pulse

repetition rate.

780 800 820 840 860 880 900 920 940 960 980 1000-5

0

5

10

15

20

25

30

35

40

45

aluminium UV-mirror reflector III reflector II reflector I

IR-r

efle

ctiv

ity

wavelength (nm)

Fig. 8: .Spectral IR-reflection index of differ-

ent materials.

clusion is, that the pulse repetition rate has no

essential influence on the efficiency at least at

small frequencies below f PULS = 25Hz.

d) The printing industry is interested in rear re-

flectors with a high reflection index in the UV-

range for great UV-efficiency and a low re-

flection index in the IR-range because of the

undesirable heat, which is produced by infra-

red radiation. Therefore five different materi-

als are tested for the reflectivity in the UV

(210-380nm)- and IR-(780-1000) wavelength

range. These five materials and their UV-re-

flection index are:

- highly polished aluminium, used so far by

Bläsing Elektronik GmbH; the UV-reflec-

tion index is ρUV = 90,24%.

- a special UV-glass-mirror that transmits the

IR-radiation and reflects the UV-radiation;

the UV-index results in ρUV = 97,17%.

- three metallic reflectors I, II and III with dif-

ferent coatings; The compositions of these

three coatings and their designation are

unkown.

During the measurements it was apparent,

that the three reflectors I, II and III are in all prob-

ability diffused and diffused/directed reflecting

materials with the consequence

that the absolute UV-reflection

index could not be determined.

But a second measurement meth-

od could prove, that all five mate-

rials could increase the radiant

power better than the aluminium

reflector.

Fig. 8 shows the spectral IR-

reflectivity of these five materials

over a wavelength range from

780-1000nm. The aluminium re-

flector is worse than the other four

materials. In comparison with all


groups and experimental results the reflector

coatings I and III are most suitable to optimize

the xenon flashlamp system.

Proposals for improvementsThe proposals to optimize the flashlamp sys-

tem can be divided in four groups:

1. gas filling

2. tube material

3. reflector material

4. reflector form

Gas fillingFour typical fillings for arc lamps are xenon

(Xe), mercury (Hg), the mixture mercury-xenon

(Hg(Xe)) and deuterium (D 2 ). To compare the

effectiveness of these fillings in the UV-area the

efficiency as a function of the input power of dif-

ferent arc lamps with these gases is calculated

over a range from 210-380nm. In Fig. 9 this com-

parison is shown. To achieve these results some

spectral irradiance curves from the L.O.T.-Oriel

company [2] catalog are evaluated. The xenon

gas lamps are plainly worse than the other

lamps. Increasing the lamp power does little in-

fluence to the UV-efficiency of the xenon burn-

D2-deuterium

Xe-xenon

Hg-mercury

Hg(Xe)-mercury/xenonη *10 9−

ηUV210 380−

lamp power (W)

Fig. 9: UV-efficiency ηUV210 380− of different discharge lamps.

er. The mixture Hg(Xe) provides the best optical

radiation power but to get the optimal power a

warm up time of about 15 minutes is necessary.

Tube materialQuartz is undoubtedly the best material for

the discharge tube and it is normally used by

the industry. Quartz guaranties the mechanical

and thermal durability. The type of quartz de-

pends on the desired UV-output. There exist

several special quartz types: a) UV grade quartz

that transmits the output to below 200nm, and

b) ozone free quartz which absorbs short wave-

lengths to prevent ozone generation. Above

280nm the special types do not offer advantag-

es compared with standard quartz variants.

Reflector materialLiterary investigations give some new infor-

mation about reflector materials to optimize the

radiation power. a) Labsphere Ltd. company

developed a diffuse reflecting material, spectral-

on, with reflectivity over the range from 250-

2500nm shown in Fig. 10. Spectralon is a ther-

moplastic resin and it is thermally

stable to > 350°C. The reflectance

is > 95% over this range. Surface

contamination only decreases the

reflectance at the lower ends of

the spectral range. B) metallic re-

flectors are very sensitive to sur-

face contamination and to over-

heating. These facts can decrease

the reflectance greatly as well as

the permanent irradiation with UV-

light.

Reflector formFig.11 shows how the spatial

distribution of radiation could be

changed when two aluminium reflectors with dif-

ferent forms are used. These forms are described

in Fig.12 and Fig.13. The influence on the distri-

bution is tested when the distance between the

tube and the reflector varies. The values are

related to the 90°-direction and to the values of a

refl

ec

tan

ce

wavelength (nm)

Fig. 10: Spectral reflection grade of spectralon [4]

0

20

40

60

80100

120

140

160

1801,51,5 1,251,0 1,01,25

angle (°)

without reflector R1 1,5 cm R1 4,0 cm R1 6,5 cm R2 1,5 cm R2 4,0 cm R2 6,5 cm

Fig. 11: Spatial distribution of the radiation

with the reflectors I/II in dependence on the

distance between tube and reflector.

3 RESEARCH88

measurement without a reflector. So you are in

position to determine the increase of the radiant

intensity as well. The reflector R2 could increase

the radiant power best, with nearly 40% com-

pared with the

values without a

reflector. The dis-

tance between

the tube and the

reflector is impor-

tant as well. The

reflector R1 pro-

duces the best

results at a dis-

tance of d=4,0cm

whereas the

second reflector

was best at a dis-

tance of

d=1,5cm.

ConclusionThe analysis of

the flashlamp system of Bläsing Elektronik GmbH

provided the following results: The flashlamp ra-

diates only for the duration of a pulse. The pulse

repetition rate up to 25Hz has no signifcant

influence on the radiation intensity. The spatial

distribution of the radiation is extremly wide. Over

an angle of 168° fifty percent of the radiant power

compared to the power in direction of 90° is still

radiated by the flashlamp.

The investigations to optimize this system

provided the following possibilties: The mixture

mercury-xenon has the best UV-efficiency and

it is suitable to use for fillings in arc lamps. The

change of the reflector form and material could

increase the radiant power as well. First exper-

imental measurments point to the assumption that

e.g. the combination of the reflector material III

with the form R2 which could both increase the

intensity best, would provide a better efficiency.

References[1] Erhardt D. Stiebner, Bruckmann´s Handbuch

der Drucktechnik, Bruckmann, München 1992

[2] L.O.T. ORIEL catalog Vol.II, Light Sources,

Monochraomators & Spectrographs, Detectors

& Detection Systems, Fiber Optics, Oriel Cor-

poration, USA, 1994

[3] Polygon flashlamps http://www.polygon1.com/

technology.html

[4] Labsphere catalog; Diffuse reflectance Coatings

And Materials, Labsphere, North Sutton, 1996

3.5.4 Construction of a flip chip de-vice for bonding integrated circuits

J. ERVENS AND R. BUß

o bond integrated circuits in flip chip tech-

nology a heating device is required, enabling

precise adjustment and soldering of the sol-

der bumps. This device was designed,

built and put into operation. Further-

more, an alternative process to electroplat-

ing [1] was tested to put solder bumps onto

microelectrodes. Therefore, testing chips

were constructed on which solder bumps

were evaporated. In the completed device test

soldering points were carried out and ana-

lyzed.

IntroductionIn the age of space-saving integration of semi-

conductor circuits the use of flip chip technolo-

gy is getting more and more interesting. A spe-

cial advantage is the possibility to connect silicon

technique to III-V-compound semiconductors,

Fig. 12: Cross-section

of the reflector R1.

Fig. 13: Cross-section

of the reflector R2.


T

which are of high importance to optoelectronic

applications.

Optoelectronics have many components with

vertical radiation. Therefore a direct connection

of the silicon substrate to the light emitting chip

is very useful for the third dimension. Another

point of interest is the self-adjustment of the

chips by the surface tension of the melted sol-

der bumps. As shown in Fig. 1 an alignment er-

ror of several micrometer in the adjustment can

be balanced out.

Description of the mode of operation of theheating device

The complete equipment consists of

- heating unit

- adjustment unit

- temperature control

- temperature measuring instrument

The heating unit is shown at the top of Fig. 2.

It comprises the halogen radiators and the mir-

ror reflectors and serves for fastening the sam-

ple holding device. In the device, which consists

of 10 millimeter thick aluminium plates, the chips

get soldered, lying on top of each other. A mem-

brane pump sucks the top chip to a heat-resis-

tant pane of glass as shown in Fig. 3. The bot-

tom chip is put down on an appliance fastened

to the adjustment unit which consists of a rotary

table, that can be moved by hand, and an x-y-z-

manipulator, see Fig. 3. The adjustment of the

chips can be observed with an ocular from the

top.

After the adjustment is complet-

ed in x- and in y-direction the

bottom chip is brought into con-

tact with the top chip by the z- ma-

nipulator. Then the nitrogen valve

is opened and a low pressure is

adjusted by a manometer, to

heat the chips in a nitrogen atmosphere. This

disables oxidation of the surfaces of the solder

bumps. The halogen radiators are started and

controlled by the temperature control unit, which

consists of a phase-angle control, that influenc-

es the electric power. By means of a rotary po-

tentiometer the temperature of the radiators is

also influenced. A fast-response thermocouple

juts as a measuring sensor into the heating unit.

This digital measuring instrument shows the pre-

dominant temperature inside. After about ten min-

utes the chips are soldered and after cooling

down thay can be taken out off the heating unit.

Fig. 2: Front view of flip chip device.

Fig 1: Self-adjustment of solder bumps [2].

3 RESEARCH90

Construction of test chipsAt first the substrate is covered with a steel

resist pattern mask and fastened in the deposi-

tion apparatus. Then a gold layer is evaporat-

ed, because the adhesive bond strength of gold

on the substrate is significantly higher than that

of soft solder. The soft solder layer is evaporat-

ed in several steps of the operation.

Results of the test seriesThe temperature in the heating unit is suffi-

cient in order to melt the solder bumps and to

connect the chips. In case of stronger mechan-

ical demand the gold layer dissolves off the sub-

strate, which means the soldering of the solder

bumps was successful. Self-adjustment can not

be recognized, because the layer thickness

achieved in the evaporation process is far too

thin. If a higher layer thickness can be achieved,

evaporation will be applicable, but regarding to-

days knowledge electroplating is preferred.

References[1] G. Sadowski, D. Zeidler: Mikrogalvanik für die

Herstellung lötfähiger Bumpsysteme, me Bd 6,

1992, pp 358-361

[2] M. Wale, M. Goodwin: Flip-Chip Bonding Opti-

mizes Opto-ICs, Circuits and Devices, pp 25-

31, Nov. 1992chip 1chip 2

xy

z

sealing ring

diminished pressure panes of glass

rotatable and inx-y-z-directionshiftable appliance

Fig. 3: Principle of chip arrangement.


4 TEACHING ACTIVITIES92

4 Teaching activities

4.1 Lectures, Excercises, andpractical studies

Technical Electronics 3: Optoelec-tronics

D. Jäger and A. Stöhr

The course Technical Electronics 3: Opto-

electronics covers the basic theory and tech-

nology of modern semiconductor photonic de-

vices as well as applications of these devices in

optoelectronic integrated circuits (OEICs). The

course starts with the fundamental physical phe-

nomenon of light-material interaction in semi-

conductors, such as fundamental absorption,

spontaneous and stimulated emission. Subse-

quent lectures deal with the theory and technol-

ogy of photoconductive devices, photodiodes,

modulators, light emitting diodes (LEDs), and

laser diodes. Special attention is given to mod-

ern quantum well waveguide laserdiodes and

their applications in optical communication sys-

tems, medicine, and material processing.

Ultra High Frequency TransmissionTechniques: Optical Signal Trans-mission

D. JÄGER AND R. BUß

The course Ultra High Frequency Transmis-

sion Techniques: Optical Signal Transmission

starts with the propagation of electromagnetic

waves considering the features of optical waves

at surface boundaries, such as reflection and

refraction. Proceeding with the description of

such fundamental physical effects like scatter-

ing, absorption and dispersion, optical wave

propagation in various types of dielectric

waveguides is discussed. Based on this funda-

mentals the design, properties and technologi-

cal realization of waveguides based on III/V com-

pound semiconductors are discussed. Another

main part of this course deals with fiber optic

waveguides: Wave propagation in graded index

fibers as well as in stepped index fibers is de-

rived where both advantages and disadvantag-

es of each type are elucidated. Problems such

as signal distortion in fibre optic waveguides are

analyzed and solutions to avoid them are giv-

en. Following the topic of wave propagation, the

most important devices for optical and optoelec-

tronic integrated circuits (OEIC) are presented.

The properties and technological realization of

waveguide laser diodes, vertical cavity surface

emitting laser diodes (VCSEL), modulators, and

detectors are discussed. Finally, economical

aspects of optical communication techniques

and future prospects like fiber to the home are

touched

Special Areas of Optoelectronics:Lasers

D. JÄGER AND A. STÖHR

The first lectures within the course Lasers

cover the basic principles and the mathemati-

cal description of electromagnetic waves. The

course proceeds with the quantum mechanical

interactions between electromagnetic waves

and atomic materials resulting in the two most

important requirements for light amplification by

stimulated emission of radiation (laser). Special

attention is then given explaining the basic con-

cepts, the functionality, and the characteristic

4.1 Lectures, excercises, and practical studies 93

specifications of different laser sources of im-

portance, such as the Helium-Neon laser, the

Ar-ion laser, Excimer lasers, the Ti:Sapphire la-

ser, semiconductor laser diodes etc.. Finally,

examples of laser applications in various indus-

trial areas (medicine, communication, material

processing etc.) are discussed together with fu-

ture trends.

Optical Signal Processing


The course Optical Signal Processing starts

with the basic theory of non-linear optical effects

both in dielectric materials and in semiconduc-

tors. The causes for optical bistability are de-

scribed and principles like optical switching are

applied to the realization of optical memories

and logic elements. Within the next section of

this course, the phenomenon of opto-electronic

bistability is introduced. It is shown that the inte-

gration of a light modulator and a photodetector

is leading to so-called self-electro-optic effect

devices (SEED), showing various forms of

switching behaviour which can be controlled

both optically and electrically. Finally, the main

advantages of optical signal processing are

pointed out while discussing applications such

as optical switching networks, image process-

ing systems, optical neural networks, optical

phased array antennas, optical computing, and

optical interconnects.

Multimedia-Techniques


This course elucidates Multimedia from

three different points of view: The optoelectron-

ic area, the informatic area and the area of data

processing. Starting with optoelectronic devic-

es and interfaces for fiber-optic networks (LAN,

WAN, FDDI), multiplexing (TDM, WDM) and

routing techniques in the optical domain are in-

troduced. Problems of high capacity data stor-

age using optical techniques and mobile con-

nections to the internet are discussed. The

second part deals with modern techniques for

data compression, coding and security problems

together with the discussion of pattern recogni-

tion using neural networks. Large electronic

databases, techniques for data retrieval, video

indexing methods and electronic data inter-

change are presented. The last part of this

course elucidates todays computer hard- and

software such as Pentium MMX technology,

multimedia PCs , WWW, Internet phone, elec-

tronic mail and more. Next, various types of net-

work protocols (ATM, FDDI, Ethernet, TCP/

IP, ...) suitable for multimedia applications are

discussed. Finally applications such as

teleteaching, teleworking, edutainment (Educa-

tion and Entertainment), video on demand, world

wide web and video conferencing are treated.

Information Technology 1 + 2

D. JÄGER AND CO-WORKERS

Practical studies for students with emphasis on

Information Technology (E3 I/IT)

Exp. 1: Optical Transmission

Exp. 2: Optical Signal Processing

Exp. 3: Optoelectronic Sensors

Exp. 4: Optical Neural Signal Processing


4.2 Seminars and Colloquia

Seminar on Optoelectronics

D. JÄGER AND CO-WORKERS

M. Groß, Dimensionierung und Entwicklung

eines thermooptischen Schalters im polynmer-

en Materialsystem, Apr. 1996

M. Alles, Tagungsberichte: IPRM 96,

22.04.-25.04., Schwäbisch Gmünd und IPR 96,

29.04.-02.05., Boston, May 1996

H. Slomka, Reinstwassererzeugung für die

Optoelektronik, May 1996

R. Hülsewede, Nichtlineare Leitungsstruk-

turen zur Frequenzerzeugung und Frequenz-

vervielfachung, May 1996

R. Buß, ZEMAX: Ein Softwarepaket zur Sim-

ulation optischer Systeme, May. 1996

M. Engel, 2D-Simulation von INT-HEMT für

OEIC, Jun. 1996

M. Wenning, Entwicklung einer Meßtechnik

zur Bestimmung der Physikalischen und fotome-

trischen Eigenschaften von LED-basierten Full-

Color-Displays, Jun. 1996

S. Redlich, Nichtlineare Vielschichthetero-

strukturen für die Microwellenphotonik, Jun.

1996

R. Hedtke, Demoskopische Untersuchung

und Beurteilung der Leserlichkeit von LED-basi-

erten Anzeigesystemen, Jun. 1996

B. Neuhaus, Aufbau einer Meßtechnik zur

Charakterisierung und Optimierung der UV-Aus-

beute von Blitzlampen für den Einsatz in der

Druckindustrie, Jul. 1996

D. Kalinowski, Entwicklung eines Feldson-

denmeßplatzes zur 2-dimensionalen Analyse

elektromagnetischer Feldverteilungen auf nich-

tlinearen Leitungen, Jul. 1996

V. Wendrix, Herstellung und Charak-

terisierung eines Wanderwellen-Photodetek-

tors, Jul. 1996

A. Kreuder, Ankopplung von Sende- und

Emfpangsmodulen an eine faseroptische Über-

tragungsstrecke, Oct. 1996

S. Redlich, Ladungsträgertransport über

Heterobarrieren - Simulationsmethoden, Oct.

1996

M. Groß, Stand des Projekts EPI-RET, Oct.

1996

A. Lüddecke, Simulation der Millimeter-

wellengeneration eines Wanderwellenphotode-

tektors, Nov. 1996

J. Evens, Aufbau einer Flip-Chip Apparatur

zur Verbindung integrierter Schaltungen, Nov.

1996

P. Karioja, Overview on activities at VTT,

Nov. 1996

4.2 Seminars and Colloquia 95

V. Mezentsev, Nonlinear Problems Related

To The Modern Optical Communication, Nov.

1996

M. Meininger, Entwicklung photovoltaischer

Zellen zur Energieversorgung einer künstlichen

Sehprothese, Dec. 1996

I. Ryjenkova, Millimeterwave propagation in

nonlinear transmission lines, Dec. 1996

O. Berger, Bestimmung des HF-Er-

satzschaltbildes von Photodetektoren mit Hilfe

der Netzwerkanalyse, Jan. 1997

T. Baumeister, Systemanalyse des

optoelektronischen Energie- und Signalübertra-

gungssystems im Projekt EPI-RET, Jan. 1997

A. Kreuder, Untersuchung der dynamischen

Eigenschaften von nichtlinearen Vielschicht-

heterostrukturen, Jan. 1997

M. Groß, Stand des Projekts EPI-RET, Feb.

1997

O. Lotz, Entwicklung einer Seelaterne in

LED-Technik in den Farben Rot und Grün, Apr.

1997

R.S. Johnson, Silicon Motherboards for Fi-

bre-Chip Coupling, Apr. 1997

M. Schmidt, Elektronische Eigenschaften

von Bor, May 1997

I. Ryjenkova, Nichtlineare Leitungen für das

Millimeterwellengebiet, Jun. 1997

T. Braasch, Bericht über die Messe Laser

1997, Jul. 1997

J. Ervens, Experimentelle Untersuchungen

zum Ladungsträgertransport über eine GaAs/

AlGaAs-Barriere, Oct. 1997

R. Heinzelmann, Bericht über die OFS97

in Williamsburg, Nov. 1997

R. Hedtke,Entwicklung einer optischen En-

ergie- und Signalübertragungsstrecke, Nov.

1997

C. Kampermann, Implementierung eines

analytischen Modells zur Simulation der optis-

chen Eigenschaften nichtlinearer Halbleiter-Het-

erostrukturen, Nov. 1997

B. Ponellis, Simulation des optischen Kon-

versionswirkungsgrades von Wanderwellen-

Photodetektoren, Dec. 1997


Colloquium on Optoelectronics

D. JÄGER AND LECTURERS WITH EMPHASIS ON OPTO-

ELECTRONICS

Prof. Dr. H.G. Schuster, Universität Kiel,

Komplexe Adative Systeme, Jan. 1996

Dipl.-Phys. G. David, Universität Duisburg,

Elektrooptische Feldverteilungsmessungen zur

Höchstfequenz-Charakterisierung von mono-

lithisch integrierten Mikrowellenschaltungen,

Feb. 1996

Dr. A.L. Ivanov, Universität Frankfurt,

Switching Kinetics of a Low-Intensity Electro-

Optical Element due to Intrinsic Photoconduc-

tivity, May 1996

Dr. J.-Uwe Meyer, Fraunhofer-Institut St. In-

gberg, Mikrotechnologien zur Kontaktierung von

biologischen Zellen und Geweben, May 1996

Dipl.-Ing. R. Heinzelmann, Universität Du-

isburg, Elektrooptische Wellenleitermodulator-

en für optische Übertragungssysteme, May

1996

Dipl.-Ing. S. van Waasen, Universität Duis-

burg, 20 Gb/s Wellenleiter-pin/Wanderwellen-

verstärker OEIC: Jüngste Ergebnisse, Jun.

1996

Ass. Prof. Dr. A. Driessen, Univ. of Ensche-

de, Netherlands, Advanced Micro-Systems for

Optical Networks (AMON), Jun. 1996

Dr. N. Vodjdani, THOMSON CSF, Orsay Ce-

dex, France, Integrated Optoelectronics for

Optical Microwave Links and Optical Communi-

cations, Oct. 1996

Prof. Dr. M. Dragoman, Time-frequency

characterization of optical pulses, Oct. 1996

Dipl.-Ing. R. Buß, Fachgebiet Optoelektro-

nik, Duisburg, Optoelektronik in der Neurotech-

nologie, Oct. 1996

Dr.-Ing. M. Martin, Hahn-Meitner-Institut,

Berlin, Entwicklung von GHz Komponenten am

Hahn-Meitner-Institut, Nov. 1996

Dipl.-Ing. M. Alles,Fachgebiet Optoelektron-

ik, Duisburg, 60 GHz Wanderwellen-Photode-

tektoren für optische Millimeterwellenverbindun-

gen, Dec. 1996

Dipl.-Phys. T. Braasch, Fachgebiet Optoele-

ktronik, Duisburg, Elektrooptisches Testen zur

on-wafer-Charakterisierung von MMICs , Jan.

1997

Dipl.-Ing. A. Brennemann, Fachgebiet Halb-

leitertechnik/-technologie, Duisburg, Neuartige

Photoreceiver auf Basis einer Kombination von

pin-Diode und Permeable Junction Base Tran-

sistor (PJBT), Jun. 1997

Prof. Dr. W. Sohler, Universität Paderborn,

Integrierte Optik in LiNbO3: neue Entwicklun-

gen, Jun. 1997

Prof. Dr.-Ing. R. Schwarte, Universität

Siegen, Neuartiges optisches 3D-Meßsystem

für die schnelle Formerfassung, Jul. 1997


4.3 Doctoral, Diploma, andGraduate theses

Doctoral theses

Gerhard David, Höchstfrequenz-Charak-

terisierung von monolithisch integrierten Mikro-

wellenbauelementen und -schaltungen durch

zweidimensionale elektrooptische Feldvertei-

lungsmessungen

Steffen Knigge, Nichtlineare Optische

Eigenschaften von Vielschichtheterostrukturen

Andreas Stöhr, Entwicklung und Real-

isierung elektrooptischer Wellenleiter-Schalter

für photonische Systeme im Wellenlängenbe-

reich um 1 µm

Ralf Kremer, Optisch gesteuerte Koplanar-

leitungen als III-V-Halbleiter-Bauelemente für die

Mikrowellen-Signalverarbeitung

Stefan Zumkley, Vertikale elektrooptische

Modulatoren für optische Verbindungstechnik im

Gbit/s-Bereich

Diploma theses

Ludger Brings, Implementierung eines

rechnergestützten Syntheseverfahrens zur Re-

alisierung monolithisch integerierter periodischer

Hochfrequenzleitungen

Peter Bussek, Quantitative Auswertung von

zweidimensionalen elektrooptischen Feldvertei-

lungsmessungen zur Charakterisierung von

monolithisch integrierten Mikrowellenschaltun-

gen

Thomas Alder, Herstellung und Charak-

terisierung von Wellenleitermodulatoren für den

Wellenlängenbereich um 1,3 µm

Michael Wenning , Entwicklung einer

Meßtechnik zur Bestimmung der physikalischen

und fotometrischen Eigenschaften von LED-

basierten Full-Color-Displays

Thomas Engel, 2D-Simulation von InP-

HEMTs für Verstärker in Empfänger-OEICs

Dirk Kalinowski, Entwicklung eines Feld-

sondenmeßplatzes zur zweidimensionalen Ana-

lyse elektromagnetischer Feldverteilungen auf

nichtlinearen Leitungen

Thomas Baumeister, Systemanalyse des

optischen Energie- und Signalübertragungs-

moduls für eine künstliche Sehprothese

Andreas Kreuder, Untersuchung der

dynamischen Eigenschaften nichtlinearer

Vielschichtheterostrukturen


Mark Meininger, Entwicklung photovoltais-

cher Zellen zur Energieversorgung einer kün-

stlichen Sehprothese (Retina Implant)

Oliver Lotz, Entwicklung einer Seelaterne

in LED-Technik in den Farben Rot und Grün

Jutta Ervens, Experimentelle Untersuchun-

gen zum Ladungsträgertransport über eine

GaAs/AlxGa1-xAs-Heterobarriere

Uwe Weimann, Entwicklung einer Infrarot-

Datenübertragungsstrecke auf rotierenden Bild-

Text-Systemen

Claus Kampermann, Implementierung

eines analytischen Modells zur Simulation der

optoelektronischen Eigenschaften nichtlinearer

Halbleiter-Heterostrukturen

4.3 Doctoral, Diploma, and Gratuate theses 101

Graduate theses

Michael Heinsdorf, Herstellung und Char-

akterisierung von Wanderwellen-Photodetek-

toren auf InP-Substrat

Ralph Hedtke, Demoskopische Untersu-

chungen zur Beurteilung der Leserlichkeit von

LED-basierten Anzeigesystemen

Birgit Neuhaus, Aufbau einer Meßtechnik

zur Charaktierisierung und Optimierung der UV-

Ausbeute von Blitzlampen für den Einsatz in der

Druckindustrie

Jutta Ervens, Aufbau einer Flip-Chip-Appa-

ratur zur Verbindung integrierter Schaltungen

André Lüdecke, Simulation der Millimeter-

wellengeneration eines Wanderwellen-Photode-

tektors

Oliver Berger, Bestimmung des HF-Er-

satzschaltbildes von Photodetektoren mit Hilfe

der Netzwerkanalyse

Bernd Ponellis, Simulation des optoelektro-

nischen Konversionswirkungsgrades von Wan-

derwellen-Photodetektoren

5 PUBLICATIONS AND PRESENTATIONS102

5 Publications and presenta-tions


Analysis of microwave propagation effects

using 2D electro-optic field mapping tech-

niques, Optical and Quantum Electronics,

Special Issue on Optical Probing of

Ultrafast Devices and Integrated Circuits,

1996, pp. 919 - 931

[2] G. David, P. Bussek, and D. Jäger, High

resolution electro-optic measurements of

2D field distributions inside MMIC devices,

Proceedings of CLEO 96, Anaheim, USA,

1996, pp. 450-451


In-circuit electro-optic field mapping for

function test and characterization of

MMICs, 1996 IEEE MTT-S Int. Microwave

Symp., June 17-24, San Francisco, USA,

1996, pp. 1533-1536

[4] R. Kremer, S. Redlich, L. Brings, and D.

Jäger, A novel type of constant impedance

travelling wave phase shifter for InP- based

MMICs, 1996 IEEE MTT-S Int. Microwave

Symp., June 17-24, San Francisco, USA,

1996

[5] M. Alles, Th. Braasch, and D. Jäger, High-

speed coplanar Schottky travelling-wave

photodetectors, Int. Conf. on Integrated

Photonics Research, Conference Proceed-

ings pp. 380-383, Boston, USA 1996

[6] G. David and D. Jäger, Analysis of in-cir-

cuit electro-optic measurements of MMICs,

XXVth General Assembly of the URSI,

August 28 -September 5, 1996, Lille,

France

[7] M. Alles, Th. Braasch, R. Heinzelmann, A.

Stöhr, and D. Jäger, Optoelectronic de-

vices for microwave and millimeterwave

optical links, 11th Int. MIKON 96, Confer-

ence Proceedings, Workshop Optoelec-

tronics in Microwave Technology, pp. 68

- 76, Warsaw, 27-30 May 1996 (invited

paper)

[8] M. Alles, R. Heinzelmann, R. Hülsewede,

R. Kremer, S. Redlich, A. Stöhr, and D.

Jäger, Wave propagation in planar struc-

tures for travelling wave semiconductor

devices, Progress In Elektromagnetic Re-

search Symposium PIERS 96, Confer-

ence Proceedings, p. 136, July 1996,

Innsbruck, Austria (invited paper)

[9] P. Berini, A. Stöhr, K. Wu, D. Jäger, Nor-

mal Mode Analysis and Characterization

of an InGaAs/GaAs MQW Field-Induced

Optical Waveguide Including Electrode Ef-

fects, IEEE/OSA J. Lightwave Technol.,

Vol. 14, No. 10, pp. 2422 - 2435, October

1996

103

[10] D. Jäger, M. Alles, T. Braasch, R.

Heinzelmann, and A. Stöhr, Integration

Technology for Microwave Photonic De-

vices, Interaction technology of Micro-

waves and Light-Waves-Systems and

Devices, XXVth General Assembly of the

URSI, August 28 -September 5, 1996, Lille,

France (invited paper)

[11] D. Jäger, R. Hülsewede, I.V. Ryjenkova,

V.K. Metzentsev, S. L. Musher, Microwave

Propagation on Nonlinear Transmission

Lines, XXVth General Assembly of the

URSI, August 28 -September 5, 1996, Lille,

France

[12] D. Jäger, V.K. Metzentsev, I.V. Ryjenkova,

S. K. Turitsyn and R. Hülsewede, Micro-

wave Propagation on Nonlinear Transmis-

sion Lines, Proceedings of PIERS96,

Hong Kong

[13] M. Alles, T. Braasch, and D. Jäger, Trav-

elling Wave Photodetector for Optical Gen-

eration of Microwave Signals, Indium

Phosphide and Related Materials IPRM

96, Proc. Part II, pp. 30 - 31, Schwäbisch

Gmünd, 1996 (post deadline paper)

[14] R. Hülsewede, V.K. Mezentsev, S.L.

Musher, I.V. Ryjenkova, S.K. Turitsyn, and

D. Jäger, Travelling wave generation of

millimeter waves in bi-modal NLTLs, 26th

European Microwave Conference EuMC

96, Prague

[15] R. Heinzelmann, A. Stöhr, Th. Alder, R.

Buß, and D. Jäger, EMC measurements

using electrooptic waveguide modulators,

International Topical Meeting on Micro-

wave Photonics MWP 96, Conference

Proceedings, Technical Digest, December

3-5, 1996, Kyoto, Japan

[16] R. Hülsewede, V.K. Mezentsev, S.L.

Musher, V. Ryjenkova, S.K. Turitsyn, and

D. Jäger, Millimeter Wave Generation on

Nonlinear Transmission Lines, 1996 Int.

Workshop on Millimeter Waves Digest,

1996, Orvieto, Italy

[17] D. Jäger, Optically Controlled Microwave

Devices, International Topical Meeting on

Microwave Photonics MWP 96 technical

digest, December 3-4, 1996, Kyoto, Japan

[18] D. Jäger, V.K. Mezentsev, S.L. Musher and

I.V. Ryjenkova, Millimeter wave power gen-

eration on nonlinear transmission lines,

Asia Pacific Microwave Conf. APMC 96,

New Delhi, India

[19] I. Ryjenkova, V.K. Mezentsev, S.L.

Musher, S.K. Turitsyn, R. Hülsewede and

D. Jäger, Nonlinear Transmission Lines for

Millimeter Wave Applications, INMMC 96,

Duisburg

[20] Th. Braasch, G. David, R. Hülsewede, U.

Auer, F.-J. Tegude and D. Jäger, Propa-

gation of Microwaves in MMICs Studied by

Time- and Frequency-Domain Electro-Op-

tic Field Mapping, Spring Topical Meeting

Ultrafast Electronics and Optoelectronics,

March 17-19, 1997, Lake Tahoe, USA


[21] Th. Braasch, G. David, R. Hülsewede, U.

Auer, F.-J. Tegude and D. Jäger, Fre-

quency and time domain characterization

of nonlinear transmission lines using

electro-optic probing techniques, MIOP

97, April 22-24, 1997, Sindelfingen, Ger-

many

[22] M. Alles, U. Auer, F.-J. Tegude and D.

Jäger, Millimeterwave Photodetectors, Mi-

crowaves and Optronics, MIOP 97, April

22-24, 1997, Sindelfingen, Germany

[23] M. Alles, U. Auer, F.-J. Tegude and D.

Jäger, High-Speed Travelling-Wave Pho-

todetectors for optical Millimeterwave

transmission operating at 1.55 µm, Work-

shop Mobile Millimeter Communications

MMMCom, Dresden, 12-13. Mai 1997

[24] S. Redlich, A. Kreuder, and D. Jäger, Dy-

namics of nonlinear electro-optical GaAs/

AlAs multilayer-heterostructures, Interna-

tional Conference on Low Dimensional

Structures (LSDS) 97, May 19-21, 1997,

Lissabon, Portugal

[25] A. Stöhr, Heterostructure Semiconductor

Photonic Devices and Systems,

Euroconference on Advanced Heterostruc-

tures, July 1997, Grenoble, France

[26] I. Ryjenkova, M. Alles and D. Jäger, Non-

linear travelling wave photodetector for mil-

limeter wave harmonic frequency genera-

tion, Journal of Communications Special

Issue, Microwave Photonics, Vol. 48, Aug.

1997, pp. 14-17

[27] I. Ryjenkova, V.K. Mezentsev, S.L.

Musher, S.K. Turitsyn, R. Hülsewede, and

D. Jäger, Millimeter Wave Generation on

Nonlinear Transmission Lines, Publication

in annales des télécommunications (spe-

cial issue), Vol. 52, No. 3-4, 1997, pp. 134-

139

[28] M. Alles, U. Auer, F.-J. Tegude, and D.

Jäger, High-speed Travelling-Wave Pho-

todetectors for Wireless Optical Millimeter

Wave Transmission,MWP 97, Sep. 3-5,

1997, Duisburg/Essen, Germany

[29] Th. Braasch, G. David, R. Hülsewede, and

D. Jäger, 1D- and 2D-elektro-optic field

mapping to study nonlinear effects in

NLTLs, MWP 97, Sep. 3-5, 1997,

Duisburg/Essen, Germany

[30] S. Redlich, and D. Jäger, Nichtlineare

Vielschichtheterostrukturen für die Mikro-

wellen-Photonik, Photonik Symposium,

Oct. 7-9, 1997, Schwäbisch Hall, Germany

[31] S. Redlich, C. Kampermann, and D. Jäger,

Vielschichtheterostrukturen: Neue

Materialien für die Mikrowellen-Photonik,

Photonik-Symposium, Oct. 8-10, 1997,

Würzburg, Germany

[32] S. Redlich, C. Kampermann, and D. Jäger,

Modeling and simulation of nonlinear hy-

brid AlGaAs/GaAs Bragg reflectors, 10th

III-V Semiconductor Device Simulation

Workshop, Oct. 16-17, 1997, Torino, Italy

105

[33] A. Stöhr, R. Heinzelmann, T. Alder, W.

Heinrich, T. Becks, D. Kalinowski, M.

Schmidt, M. Groß, and D. Jäger, Optically

Powered Integrated Optical E-Field Sen-

sor, 12th International Conference on Op-

tical Fiber Sensors, Conference Proceed-

ings, Oct. 1997, Williamsburg, Virginia,

USA

[34] M. Groß, T. Alder, R. Buß, R. Heinzelmann,

M. Meininger, and D. Jäger, Micro Photo-

voltaic Cell Array for Energy Transmission

into the Human Eye, EPVSEC14, 1997,

Barcelona, Spain, Vol. 1, pp. 1165 - 1167

[35] M. Alles, U. Auer, F.-J. Tegude, and D.

Jäger, High-Speed Travelling-Wave Pho-

todetectors for Optical Generation of

Millimeterwaves, APMC 97, Dec. 2-5,

1997, Hongkong, China

[36] I. Ryjenkova, D. Jäger, Nonlinear RTD Cir-

cuits for High-Speed A/D Conversion,

APMC 97, Dec. 2-5, 1997, Hongkong,

China


107

6 Guide to the Department of Optoelectronics

Travel by car - The Department of Optoelectronics, now located in the Center for Solid-State Elec-

tronics and Optoelectronics (ZHO), can easily reached by car via various highways: A3 from the

north and south, A40 from the east and west. Exit at Duisburg-Kaiserberg or Duisburg-Wedau, see

map for details.

Travel by plane - From Düsseldorf International Airport take the city-train (S-Bahn) S1 to Duis-

burg main station (Hauptbahnhof, Hbf).

Travel by train - From Duisburg main station (Hauptbahnhof, Hbf) it is a 20 min. walk to the Depart-

ment of Optoelectronics and the ZHO. You can either go by Taxi or take the bus 933 or 936 to

Universität or take the tram 901 to station Zoo/Uni.

108

Notes:

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